Method, system, and apparatus for remote data calibration of a RFID tag population

ABSTRACT

A method, system, and apparatus for remotely calibrating data symbols received by a radio frequency identification (RFID) tag population are described. Tags are interrogated by a reader, which may be located in a network of readers. The reader transmits data symbols to the tags. Tags respond to the interrogations with symbols that each represent one or more bits of data. To calibrate the tags, the reader transmits a plurality of pulses of different lengths to the tag population. The tags receive the plurality of pulses. A characteristic of each pulse, such as a pulse length, is stored by the tags. The stored pulse lengths are used to define different data symbols that are subsequently received by the tags from the reader.

This application is a continuation of U.S. application Ser. No.10/926,269, filed Aug. 26, 2004, is now a U.S. Pat. No. 6,956,509, whichis a continuation of U.S. application No. 10/072,870, filed Feb. 12,2002, now U.S. Pat. No. 6,784,813, which claims the benefit of U.S.Provisional Application No. 60/267,713, filed Feb. 12, 2001, which areall herein incorporated by reference in their entirety.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates generally to radio frequencyidentification (RFID) tags.

2. Description of the Related Art

Many product-related and service-related industries entail the useand/or sale of large numbers of useful items. In such industries, it maybe advantageous to have the ability to monitor the items that arelocated within a particular range. For example, within a particularstore, it may be desirable to determine the presence of inventory itemslocated on the shelf, and that are otherwise located in the store.

A device known as an RFID “tag” may be affixed to each item that is tobe monitored. The presence of a tag, and therefore the presence of theitem to which the tag is affixed, may be checked and monitored bydevices known as “readers.” A reader may monitor the existence andlocation of the items having tags affixed thereto through one or morewired or wireless interrogations. Typically, each tag has a uniqueidentification number that the reader uses to identify the particulartag and item.

Currently available tags and readers have many disadvantages. Forinstance, currently available tags are relatively expensive. Becauselarge numbers of items may need to be monitored, many tags may berequired to track the items. Hence, the cost of each individual tagneeds to be minimized. Furthermore, currently available tags consumelarge amounts of power. Currently available tag power schemes, whichinclude individually tag-included batteries, are inefficient andexpensive. These inefficient power schemes also lead to reduced rangesover which readers may communicate with tags in a wireless fashion.Still further, currently available readers and tags use inefficientinterrogation protocols. These inefficient protocols slow the rate atwhich a large number of tags may be interrogated.

Hence, what is needed is a tag that is inexpensive, small, and hasreduced power requirements. Furthermore, what is needed are moreefficient tag interrogation techniques, that operate across longerranges, so that greater numbers of tags may be interrogated at fasterrates.

SUMMARY OF THE INVENTION

The present invention is directed to an RFID architecture where a readeror network of readers may interrogate and/or power tags at data rates,distances, and reliability levels that are greater than those currentlyattainable in the RFID industry for similarly classified tags. Thepresent invention is also directed to RFID tags that may be produced atcosts lower than those of similarly classified tags that are currentlyavailable. These features are the result of a unified design approach,where the attainment of higher data rates and increased communicationdistances are correlated with the realization of a lower tag cost.

In particular, the tag of the present invention has the advantages oflower cost and reduced power consumption. The reduction of powerconsumption in turn increases the possible communication range between areader and a tag. The connection between the reduction of cost and powerconsumption is based at least in part on the principle that the powerconsumption of an electronic device is directly proportional to (1) thenumber of transistors in the device, and (2) the frequency at whichcircuits of the device operate. The tags of the present inventionoperate according to efficient algorithms, such as binary traversalcommunications protocols, that require reduced logic processing. Thereduction in logic processing promotes the use of fewer transistors.Furthermore, the use of fewer transistors promotes reduced powerconsumption and reduced circuit sizes. Reduced circuit sizes lower thecost of producing the one or more chips that host the circuits, whichmay be application specific integrated circuits (ASIC), for example.

The present invention utilizes the reduction in power consumption toincrease the range at which a reader and a tag may communicate. Tags ofthe present invention may incorporate charge pump circuitry and anenergy storage capacitor to convert RF energy received from readers intoan operational voltage and current. This conversion occurs at a greaterdistance from the reader than currently existing tags allow. By reducingthe tag's power consumption requirements, the present invention enablesa tag to communicate with a reader across greater distances, where themagnitude of RF energy received from the reader is insufficient to powerconventionally designed tags.

The high data rates provided by the present invention are the result ofsimplified algorithms. Conventional algorithms interrogate RFID tagswith highly complex algorithms that are less efficient. Furthermore, theuse of conventional algorithms often involves two or more RFtransmissions colliding. When collisions occur, further RF transmissionsare required to resolve the collisions. Collision resolution consumestime without conveying data, and mandates complexity in algorithms,circuit design, and transistor count. In contrast, the present inventionemploys simple, efficient algorithms, such as binary traversalprotocols, that enable a population of tags to be interrogated in acollision free environment.

This collision free environment is based on a set of communicationssymbols that allows multiple symbol values to be transmittedsimultaneously without interference between, or destruction of any ofthese symbols.

These simplified algorithms do not require complex circuitry and intenseprocessing. Therefore, in addition to promoting increased data rates,these efficient algorithms also promote lower transistor counts, lowertag costs, and lower power consumption (which increases the range atwhich a reader and tag may communicate).

A method, system, and apparatus for defining data signal symbols in aRFID tag device are described herein. A first calibration pulse isreceived on an input signal. A length of the received first calibrationpulse is stored as a stored first length. A second calibration pulse isreceived on the input signal. A length of the received secondcalibration pulse is stored as a stored second length.

Data symbols are detected according to the present invention. A datasymbol having a pulse portion is received on the input signal. The pulseportion has a length. The data symbol is determined to be a first datavalue if the length of the pulse portion is less than the stored firstlength. The data symbol is determined to be a second data value if thelength of the pulse portion is greater than the stored first length andless than the stored second length. The data symbol is determined to bea third data value if the length of the pulse portion is greater thanthe stored second length.

In a further aspect of the present invention, data symbols in the RFIDtag device may be further defined. A third calibration pulse is receivedon the input signal. A length of the received third calibration pulse isstored as a stored third length.

Data symbols are detected according to this further aspect of thepresent invention. A data symbol is received having a pulse portion onthe input signal. The pulse portion has a length. The data symbol isdetermined to be a first data value if the length of the pulse portionis less than the stored first length. The data symbol is determined tobe a second data value if the length of the pulse portion is greaterthan the stored first length and less than the stored second length. Thedata symbol is determined to be a third data value if the length of thepulse portion is greater than the stored second length and less than thestored third length.

The first, second, and third data values may be defined to represent anyvariety of data values. In one example aspect of the present invention,the first data value is defined as a 0 bit, the second data value isdefined as a 1 bit, and the third data value is defined as a Null bit.

BRIEF DESCRIPTION OF THE DRAWINGS

The present invention will be described with reference to theaccompanying drawings. In the drawings, like reference numbers generallyindicate identical, functionally similar, and/or structurally similarelements. The drawing in which an element first appears is indicated bythe leftmost digit(s) in the reference number.

FIG. 1 is a block diagram of an environment where an RFID tag readernetwork communicates with one or more RFID tags, according to anembodiment of the present invention.

FIG. 2 is a block diagram illustrating an architectural overview ofcommunications between a reader network and a tag, according to anembodiment of the present invention.

FIGS. 3–5 are plots of example data symbols transmitted by a reader,according to embodiments of the present invention.

FIGS. 6–9 are plots of example backscatter symbols sent from a tag to areader, according to embodiments of the present invention.

FIGS. 10 and 11 are block diagrams illustrating functionalimplementations of RFID tags, according to embodiments of the presentinvention.

FIG. 12A is a state diagram illustrating various operating states of anRFID tag, according to an embodiment of the present invention.

FIGS. 12B–12D are signal representations of master reset and masterdormant signal conditions, according to embodiments of the presentinvention.

FIG. 13 is a flowchart that illustrates an operation of a binarytraversal protocol from the perspective of a single tag, according to anembodiment of the present invention.

FIGS. 14A and 14B are flowcharts that illustrate example specific readinterrogation operations from the perspective of a reader, according toembodiments of the present invention.

FIGS. 15A and 15B are flowcharts illustrating example general readinterrogation operations from the perspective of a reader, according toembodiments of the present invention.

FIG. 16 illustrates a example tree diagram describing the binarytraversal of a population of three tags, according to an embodiment ofthe present invention.

FIG. 17A is a flowchart that illustrates an operation of a superpositionsubset of the protocol from the perspective of a single tag, accordingto an embodiment of the present invention.

FIG. 17B is a flowchart that illustrates an operation of a superpositionsubset of the protocol from the perspective of a reader network,according to an embodiment of the present invention.

FIG. 18 illustrates a frequency selectable oscillator for use in adigital synchronous circuit driven by a master clock signal.

FIG. 19 shows an oscillator configuration that provides for multiplesimultaneous oscillator frequencies, according to an embodiment of thepresent invention.

FIG. 20 illustrates an example block diagram of a frequency adjustableoscillator with tuning circuits, according to an embodiment of thepresent invention.

FIG. 21A shows an oscillator calibration circuit, according to anembodiment of the present invention.

FIGS. 21B and 21C illustrate a more detailed block diagram of thecalibration circuit of FIG. 21A, according to an embodiment of thepresent invention.

FIG. 21D illustrates a more detailed of a frequency adjustment bank,according to an embodiment of the present invention.

FIG. 22A shows an example waveform used for one of a series ofcalibration tests, according to an embodiment of the present invention.

FIG. 22B shows an example series of test waveforms used for a fullcalibration, according to an embodiment of the present invention.

FIG. 23A shows an example value for a count word, according to anembodiment of the present invention.

FIG. 23B shows an example value for a control word, according to anembodiment of the present invention.

FIG. 24 shows a block diagram for an example successive approximationregister, according to an embodiment of the present invention.

FIGS. 25A–25D show flowcharts providing steps for calibrating anoscillator frequency with an input signal, according to embodiments ofthe present invention.

FIG. 26A illustrates example waveforms that may be received by a tag tocalibrate data symbols, according to an embodiment of the presentinvention.

FIG. 26B is a full data symbol timing chart depicting interactionbetween RFID readers and tags on each symbol exchange according toembodiments of the present invention.

FIG. 27 shows a data calibration system in a tag, according to anembodiment of the present invention.

FIGS. 28A–28F show flowcharts providing steps for performing data symbolcalibration and interpreting received data symbols, according toembodiments of the present invention.

FIG. 29 shows a test waveform with additional spacing pulses that may beused instead of the test waveform shown in FIG. 22A to calibrate anadjustable oscillator, according to an embodiment of the presentinvention.

FIG. 30 illustrates an analog front-end of an exemplary RF tag,according to an embodiment of the present invention.

FIG. 31 illustrates a power charge pump that is an embodiment of themain charge pump of the analog front-end shown in FIG. 30.

FIGS. 32A–32C illustrate diode curves associated with the diodes in thepower charge pump 3100.

FIGS. 33A and 33B illustrate a DC output voltage and charge pumpefficiency verses the RF input power when using diodes to limit theoutput voltage of the power charge pump 3100.

FIGS. 34A and 34B further illustrate charge pumps, according toembodiments of the present invention, where the diodes in each stage arereplaced with metal oxide field effect transistors (MOSFET) that areconfigured as diode equivalents devices.

FIGS. 35A and 35B illustrate an unbiased MOSFET having a gate terminal,a drain terminal, a body terminal, and a source terminal.

FIGS. 36A–36C illustrate a MOSFET biased as a conventional load device.

FIGS. 37A–37C illustrate a MOSFET diode biased according to the presentinvention so as to lower the threshold voltage of MOSFET diodeconfiguration and to prevent reverse bias conduction.

FIGS. 38A–38C illustrate a comparison of the IV curve for the MOSFETdiode with the IV curve of a conventional MOSFET device, and with the IVcurve of a Schottky diode.

FIG. 39 illustrates the effect of lowering the threshold voltage for aconventional MOSFET by adjusting the doping levels.

FIGS. 40A and 40B illustrate charge pumps according to furtherembodiments of the present invention, where the gate of one MOSFET diodeis forward biased with the output of another MOSFET diode.

FIGS. 41A and 41B illustrate exemplary IV curves that illustrate forwardbiasing of a diode.

FIG. 42 illustrates a data recovery circuit that is an embodiment of thedata recovery circuit shown in FIG. 30, according to the presentinvention.

FIG. 43A illustrates an example RF signal that is amplitude modulated.

FIG. 43B shows how a reference voltage at a node generally follows andapproaches the demodulated output signal.

FIG. 44 illustrates a fast charge pump, according to an embodiment ofthe present invention.

FIG. 45 further illustrates the backscatter switch shown in FIG. 30.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

1. Architectural Embodiments of the Present Invention

1.1 Tag Interrogation Environment

Before describing the invention in detail, it is helpful to describe anexample environment in which the invention may be implemented. Thepresent invention is particularly useful in radio frequencyidentification (RFID) applications. FIG. 1 illustrates an environmentwhere an RFID tag reader network 104 communicates with an exemplarypopulation of RFID tags 120, according to the present invention. Asshown in FIG. 1, the population of tags 120 includes a first tag 102 a,a second tag 102 b, a third tag 102 c, a fourth tag 102 d, a fifth tag102 e, a sixth tag 102 f, and a seventh tag 102 g. These seven tags 102are shown in the population of tags 120 for exemplary purposes.According to embodiments of the present invention, a population of tags120 may include any number of one or more tags 102. In some embodiments,very large numbers of tags 102 may be included in a population of tags120, including hundreds, thousands, or even more tags 102.

As shown in FIG. 1, one or more interrogation signals 110 aretransmitted from reader network 104 to the population of tags 120. Oneor more response signals 112 are transmitted from RFID tags 102 toreader network 104. For example, as shown in FIG. 1, first tag 102 atransmits a first response signal 112 a, second tag 102 b transmits asecond response signal 112 b, third tag 102 c transmits a third responsesignal 112 c, fourth tag 102 d transmits a fourth response signal 112 d,fifth tag 102 e transmits a fifth response signal 112 e, sixth tag 102 ftransmits a sixth response signal 112 f, and seventh tag 102 g transmitsa seventh response signal 112 g.

According to the present invention, signals 110 and 112 are exchangedbetween reader network 104 and tags 102 according to one or moreinterrogation protocols. An exemplary protocol is the binary traversalprotocol that is described below. The binary traversal protocol, incombination with other features of the present invention as describedherein, efficiently avoids collisions between signals transmitted bytags 102 so that communications bandwidth is conserved and interrogationtimes are minimized. However, other interrogation protocols may beemployed. Examples of such alternative protocols are described in U.S.Pat. No. 6,002,344 issued Dec. 14, 1999 to Bandy et al., entitled“System and Method for Electronic Inventory,” which is incorporatedherein by reference in its entirety.

Signals 110 and 112 are wireless signals, such as radio frequency (RF)transmissions. Upon receiving a signal 110, a tag 102 may produce aresponding signal 112 by alternatively reflecting and absorbing portionsof signal 110 according to a time-based pattern. The time-based patternis determined according to information that is designated fortransmission to reader network 104. This technique of alternativelyabsorbing and reflecting signal 110 is referred to herein as backscattermodulation. Tags 102 may employ various approaches to performbackscatter modulation. In one such approach, tags 102 vary theimpedance characteristics of onboard receive circuitry, such as one ormore antennas and/or other connected electronic components.

Each tag 102 has an identification number. In certain embodiments, eachof tags 102 has a unique identification number. However, in otherembodiments, multiple tags 102 may share the same identification number,or a portion thereof. During the aforementioned communications with tags102, reader network 104 receives identification numbers from tags 102 inresponse signals 112. Depending on the protocol employed for suchcommunications, the retrieval of identification numbers from tags 102may involve the exchange of signals over multiple iterations. In otherwords, the receipt of a single identification number may require readernetwork 104 to transmit multiple signals 110. In a corresponding manner,tags 102 will respond with respective signals 112 upon the receipt ofeach signal 110, if a response is appropriate.

Alternatively or in addition to identification numbers, reader network104 may send other information to tags 102. For example, reader network104 may store a unit of information in one or more of tags 102 to beretrieved at a later time. Depending upon the design of tags 102, thiscould be volatile or non-volatile information storage and retrieval.

Reader network 104 may also obtain information generated by sensors thatare included in tags 102. When provided to reader network 104, thissensor information may include information regarding the operationalenvironments of tags 102, for example.

A variety of sensors may be integrated with tags 102. Exemplary sensorsinclude: gas sensors that detect the presence of chemicals associatedwith drugs or precursor chemicals of explosives such as methane,temperature sensors that generate information indicating ambienttemperature, accelerometers that generate information indicating tagmovement and vibration, optical sensors that detect the presence (orabsence) of light, pressure sensors that detect various types oftag-encountered mechanical pressures, tamper sensors that detect effortsto destroy tags and/or remove tags from affixed items, electromagneticfield sensors, radiation sensors, and biochemical sensors. However, thislist is not exclusive. In fact, tags 102 may include other types ofsensors, as would be apparent to persons skilled in the relevant arts.

Each of tags 102 is implemented so that it may be affixed to a varietyof items. For example a tag 102 may be affixed to airline baggage,retail inventory, warehouse inventory, automobiles, and other objects.An exemplary tag implementation is described below with reference toFIG. 10.

Thus, reader network 104 may monitor the existence of, and the locationof items having tags affixed thereto, through one or more interrogationsusing the protocols referenced herein.

FIG. 2 is a block diagram of an example reader architecture 200providing communications between reader network 104 and tags 102,according to an embodiment of the present invention. Reader architecture200 includes a user application domain 230, a reader network 104, andone or more tags 102. These components are described in further detailas follows. Note that the invention is applicable to a single readerthat is communicating with tags 102, as well as to a plurality of readercoupled in a network, as in reader network 104 shown in FIG. 2. Hence,although “reader network 104” is often referred to herein, it should beunderstood that the present invention is applicable to any number ofreaders, including a single reader and multiple readers coupled in anetwork, as is required by a particular application.

At a high level, reader network 104 receives requests regarding one ormore of tags 102 from user application domain 230. Reader network 104communicates with one or more of tags 102 regarding the requests via aprotocol 214. In other words, reader network 104 transmits one or morerequests 110 to tags 102, and tags 102 respond by transmitting one ormore responses 112 to reader network 110, using protocol 214. Protocol214 is typically one of the binary traversal protocols further describedelsewhere herein.

User application domain 230 may include any number of one or more userapplications. For example, user applications of user application domain230 may include host systems such as personal computers, servers,hand-held processing devices, cell phones, and other wired or wirelessnetwork accessible devices. In the embodiment of FIG. 2, userapplication domain 230 includes wide area network (WAN) applications 202a (remote) and local applications 202 b (local). WAN and localapplications 202 a and 202 b each include user applications that usereader network 104 to access one or more of tags 102.

User applications on any number of one or more networks may communicatewith reader network 104. The networks may be of an industry standardformat and/or may be non-standard networks. In FIG. 2, applications 202a and 202 b are respectively coupled to a WAN 203 a and a local areanetwork (LAN) 203 b. In the embodiment of FIG. 2, WAN 203 a and LAN 203b are coupled together, but in alternative embodiments, they may beisolated. In the embodiment shown in FIG. 2, LAN 203 b services thephysical connection from both of applications 202 a and 202 b to readernetwork 104.

Reader network 104 includes one or more sensor interface modules (SIMs)and a remote access sensor module (RASM) domain 240. In the embodimentof FIG. 2, reader network 104 includes first and second SIMs 204 a and204 b. RASM domain 240 includes one or more RASMs 206.

Any number of one or more SIMs in reader network 104 may be used tocouple external networks to reader network 104. Accordingly, each SIMincludes an applicable hardware and/or software interface for couplingthe networks. As shown in FIG. 2, first SIM 204 a couples LAN 203 b toreader network 104.

Each SIM 204 connects to one or more RASMs 206 in reader network 104 viaone or more RASM network connections. RASMs 206 are readers that eachinclude hardware and/or software as necessary to interface with a RASMnetwork connection. As shown in FIG. 2, a first RASM network 205 acouples SIM 204 a to a number “n” of RASMs 206, and a network 205 bcouples SIM 204 a to a number “m” of RASMs 206. Such networks may carryonly data, as in network 205 b, or may carry data and power, as innetwork 205 a. One or more wiring blocks, such as wiring blocks 244 a,244 b, 244 c, 244 d, 244 e, and 244 f may be used to provide aconnection point from networks 205 a and 205 b to a respective RASM. Awiring block 244 may be an industry standard wiring block, ornon-standard type. A modified wiring block, shown as wiring block 242,may be used to inject power onto a wire or cable of one of more ofnetworks 205 a and 205 b. Operating power is provided to the RASMs 206by one or more power supplies, such as by power supplies 246, 248, 250,and 252, which may or may not be the same device.

A RASM 206 communicates with a tag 102 via one or more antenna(s) 210.Accordingly, each RASM 206 includes one or more transmitters andreceivers that are coupled antennas 210. The transmitters and receiversmay be of any variety of types. A variety of antenna configurations areavailable. In an embodiment, RASM 206 a, which is coupled to network 205a, is directly connected to up to four antennas (e.g., antennas 210a–210 d, shown in FIG. 2). In an alternative embodiment, a RASM iscoupled to and controls a multiplexer. For example, as shown in FIG. 2,RASM 206 a, which is coupled to network 205 b, is couple to multiplexer208. A multiplexer allows for a greater number of antennas to beswitched to a single antenna port of RASM 206 b. As shown in FIG. 2,multiplexer 208 connects a single antenna port of RASM 206 b to eightantennas (e.g., antennas 211 a–211 h, shown in FIG. 2). In this manner,RASM 206 b may accommodate up to 32 antennas. Such a configuration wouldrequire three additional multiplexers 208 to be connected to antennaports of RASM 206 b. RASMs 206 are able to communicate to RFID tags 102via radio frequencies using one or more protocols 214.

Furthermore, in an embodiment, each RASM 206 includes logic thatdetermines data values for symbols received from one or more tags 102that are modulated according to backscatter modulation techniquesdescribed herein. For example, the logic may determine that a receivedbackscatter symbol represents a first logical value (i.e., a “0” or a“1” bit) when the backscatter frequency of the received backscattersymbol is determined to include a first frequency, and determines thatthe received backscatter symbol represents a second logical value whenthe backscatter frequency of the received backscatter symbol isdetermined to include a second frequency. In alternative embodiments,the functions of the logic may be incorporated in other components ofreader network 104.

An object of reader architecture 200 is to provide a reasonably pricedRFID system to the commercial marketplace in large volumes. In anembodiment, the architecture performs a batch type of operation. In abatch type of operation, a reader network 104 scans item-level assets atentry and exit points in a given space. This provides information as toa history of the asset, but does not provide confirmed information aboutthe actual asset at the time of inquiry (real time information). Devicesoperating according to the present invention are designed such that asan industry converts to real time mode, the space to be controlledbecomes covered by a reader network 104/antenna 210, 211 for real timeinformation. In such an environment, there may be many RASMs206/antennas 210,211 at the entrance and exit points to a given space.Thus, a design goal may be to reduce cost as much as possible at theRASM 206 level. According to the present invention one or more functionsare removed from the RASM 206 level, and included in the relativelysmaller number of one or more SIMs 204. The SIMs 204 exist at a higherarchitecture level, and their cost may be amortized across large numbersof RASMs.

Additionally, devices according to the present invention are required tobe compatible with legacy systems and applications at a very higharchitecture level. This allows these devices to be standardized on oneversion for all uses. However, this may also introduce a burden on theimplementation to add functionality, memory, processing power, etc., inorder to present information to the highest OSI model layers. Currently,the industry stipulates that a reader include an XML presentation layerconnection (item 203 b), with substantial buffering and filteringcapabilities. Conventional reader products attempt to add thisfunctionality into each reader. However, the approach of the presentinvention is to add this functionality only once at a gateway device,SIM 204, so that the cost of implementation is not multiplied by eachRASM 206 read point.

Hence, in embodiments of the present invention, RASM 206 is responsiblefor converting digital network requests into RF signals forcommunication with RFID tags 102. To further reduce cost, in anembodiment, each RASM 206 is configured to handle up to 4 input/outputports for antennas 210, 211, which are also referred to as “readpoints.” The cost per read point is reduced to about 25% of the cost ofsingle read point devices. In other embodiments, more or fewer antennasmay be present as required by the specific application and costs.

Furthermore, in embodiments of the present invention, SIM 204 is coupledto, and controls a plurality of RASMs. For example, a single SIM 204 maybe coupled to 50 RASMs 206. This may be done through multiple RASMnetworks 205 a and 205 b coupled to SIM 204 a. SIM 204 implements thehigh-level protocol visibility layer, adds reasonable buffers, andimplements logic to filter out any undesirable conditions for a givenapplication. SIM 206 is coupled to industry standard networking asdepicted in item 203 b (e.g., Ethernet & TCP/IP), and connects to localand remote applications in their native level, such as XML.

1.2 Wireless Interface (Protocol Domain)

1.2.1 Reader Transmitted Signals

In an embodiment, reader network 104 transmits signals, such as signal110, to tags 102 as amplitude modulated (AM) signals. For example, thetransmitted signals may be narrowband AM signals. According to thisapproach, reader network 104 varies the amplitude of a carrier signalover a specific period of time that is a function of the informationthat it is transmitting. In alternative embodiments, other modulationschemes known by persons skilled in the relevant arts may be used byreader network 104 to communicate with tags 102.

Reader network 104 conveys this information in the form of one or moresymbols that are each selected from a symbol set. FIGS. 3, 4, and 5 eachillustrate a plot of a symbol of an exemplary symbol set that includesthree symbols. In particular, FIG. 3 illustrates a plot of a symbol 302that represents a logical “0,” FIG. 4 illustrates a plot of a symbol 402that represents a logical “1,” and FIG. 5 illustrates a plot of a symbol502 that represents a “NULL” symbol. The “NULL” symbol may be used inperformance of certain calibration procedures, as well as to affect orreset the operational states of tags 102. Further details regarding theuse of “NULL” symbols are provided below.

For each of symbols 302, 402, and 502, reader network 104 varies theamplitude of a transmitted carrier signal between two values. Thesevalues are shown in FIGS. 3, 4, and 5 as S_(high) and S_(low). Thisvariation in amplitude between S_(high) and S_(low) occurs over anamount of time that is referred to herein as a symbol exchange period,T_(S). FIGS. 3, 4, and 5 show T_(S) being 12.5 microseconds. However,embodiments of the present invention may employ other values of T_(S),which may be provided either statically or dynamically (i.e., “on thefly”).

The beginning of each symbol exchange period is referred to herein as aclock start time, T_(CS). The clock start time designates when readernetwork 104 changes the amplitude of its carrier signal from S_(high) toS_(low) (referred to herein as a “falling edge”). Thus, T_(CS) signifiesthe beginning of a period of time when the carrier signal amplitude isS_(low). This period of time ends when reader network 104 changes thevalue of the carrier amplitude from S_(low) to S_(high) (referred toherein as a “rising edge”). For the symbol set of symbols 302, 402, and502, reader network 104 designates the duration of this time periodaccording to the symbol that is being transmitted.

For instance, FIG. 3 shows that when transmitting a logical “0” symbol302, reader network 104 maintains its carrier signal amplitude atS_(low) for a time duration of T_(A). However, when transmitting alogical “1” symbol 402, FIG. 4 shows that reader network 104 maintainsthe carrier amplitude at S_(low) for a time duration of T_(B). FIG. 5shows that when transmitting a “NULL” symbol 502, reader network 104maintains the carrier amplitude at S_(low) for a time duration of T_(C).Exemplary values for T_(A), T_(B), and T_(C) are 3.0 microseconds, 6.0microseconds, and 9.5 microseconds, respectively. However, the use ofother values is within the scope of the present invention.

According to the present invention, various amplitude levels forS_(high) and S_(low) may be employed. For example, In oneimplementation, S_(low) is 70% of S_(high). In other words, S_(low) isnot necessarily a 0 V amplitude signal, but can have other amplitudevalues. This provides reader network 104 with the capability to providetags 102 with more RF energy at times when it is transmitting itscarrier signal at S_(low) than a 0% S_(low) implementation (i.e.,strictly on/off keying). The invention is also applicable to otherrelative percentages for S_(high) and S_(low), including a 0% S_(low)implementation.

Tags 102 employ various timing parameters to decode symbols transmittedby reader network 104. To aid in decoding the symbol set shown in FIGS.3, 4, and 5, each tag 102 employs three timing parameters (also referredto herein as timing points) that are referred to herein as timing pointsT0, T1, and T2. Examples of timing points T0, T1, and T2 are shown inFIG. 26B. In a preferred embodiment, timing points T0, T1, and T2 areprovided to tag 102 by reader network 104 during a process referred toas data calibration, as described below.

T0 and T1 correspond to points in time after the clock start time,T_(CS), that are used to distinguish between different symbol values. Inparticular, T0 is set by reader network 104 to the midpoint of theelapsed time before the rising edge associated with a logical “0” symbol302 (i.e., T_(A)) and the elapsed time associated with the rising edgeof a logical “1” symbol 402 (i.e., T_(B)). T1 is set by reader network104 to the midpoint of the elapsed time before the rising edge of alogical “1” symbol 402 (i.e., T_(B)) and the elapsed time before therising edge of a “NULL” symbol 502 (i.e., T_(C)). In an embodiment, T2corresponds to the moment in time where tags 102 need to stop theirtransmissions and return to a listening state to reader network 104. T2is preferably set by reader network 104 to a point in time slightlybefore the next T_(CS) from the reader.

In an embodiment, tag 102 employs these time parameters, which may besent by the reader during a calibration sequence, to determine theidentity of a data symbol received from reader network 104 in thefollowing manner: First, tag 102 initializes a counter or timer upon theoccurrence of a falling edge on a received signal. This initializationcoincides with a T_(CS) for a transmitted symbol. Next, the timerincrements with the passage of time until tag 102 detects a rising edgein the received signal. After the rising edge is detected, the tag 102performs a comparison between the timer value and the timing points.Namely, tag 102 detects a logical “0” symbol 302 when a rising edgeoccurs before the timer reaches T0. However, if a rising edge occurs onor after the timer reaches T0, but before it reaches T1, tag 102 detectsa logical “1” symbol 402. Alternatively, if a rising edge occurs on orafter it reaches T1, tag 102 detects a “NULL” symbol 502. This approachdynamically accommodates variations in timing between different tags 102that would cause communication errors in other more exacting timingschemes.

1.2.2 Tag Transmitted Signals

As described above, tags 102 may send information to reader network 104in the form of backscatter modulated signals. Backscatter modulationrefers to the technique of alternatively absorbing and reflecting thesignal transmitted by reader network 104. These backscatter modulatedsignals may convey symbols that are each transmitted in response to acorresponding symbol transmitted by reader network 104. Thus, each tag102 may transmit one or more backscatter symbols that are each selectedfrom a backscatter symbol set. An example of a backscatter symbol set isdescribed herein with reference to FIGS. 6, 7, 8, and 9. This symbol setuses two frequencies as a basis for sub-modulating backscatter energy.One frequency is used to transmit a logical “0” bit, while the otherfrequency is used to transmit a logical “1” bit. Note that inalternative embodiments, two different phase delays, two differentsignal amplitudes, or a single frequency or phase delay used during twodifferent time periods, may also be used to represent different logicalbit values according to backscatter modulation techniques.

The backscatter symbol set that is shown in FIGS. 6, 7, 8, and 9operates with the reader transmitted symbol set described above withreference to FIGS. 3, 4, and 5. In particular, this backscatter symbolset provides for the modulation of the latter portion of these readertransmitted symbols. As described above, these latter portions have amagnitude of S_(high).

FIGS. 6 and 7 illustrate backscatter symbols that each represent alogical “0” bit transmitted from tag 102 in modulated backscatter. Inparticular, FIG. 6 illustrates a backscatter transmitted logical “0”symbol 602 from tag 102 responding to a reader-originated logical “0”symbol 302. FIG. 7 illustrates a backscatter transmitted logical “0”symbol 702 from tag 102 responding to a reader-originated logical “1”symbol 402. Each of these symbols includes a series of pulses occurringat a certain frequency. As shown in FIGS. 6 and 7, the pulses for eachof these backscatter symbols 602 and 702 continue until the end of thesymbol exchange period, T_(S). However, each of backscatter symbols 602and 702 starts at a distinct time.

These distinct start times occur because the reader transmitted “0” and“1” symbols 302 and 402, as described above with reference to FIGS. 3and 4, each have a distinct rising edge time. Namely, the rising edgeassociated with a reader-originated “0” symbol 302 occurs at T_(A)(e.g., 3 microseconds), while the rising edge associated with areader-originated “1” symbol 402 occurs at T_(B) (e.g., 6 microseconds).

FIGS. 8 and 9 illustrate symbols that each represent a logical “1” bittransmitted from tag 102 in modulated backscatter. In particular, FIG. 8illustrates a backscatter transmitted logical “1” symbol 802 from tag102, which is responding to a reader-originated logical “0” symbol 302.In contrast, FIG. 9 illustrates a backscatter transmitted logical “1”symbol 902 from tag 102, which is responding to a reader-originatedlogical “1” symbol 402. Each of backscatter symbols 802 and 902 includesa series of pulses occurring until the end of the symbol exchangeperiod, T_(S). These pulses repeat at a frequency that is different thanthe frequency used for the logical “0” backscatter symbols 602 and 702of FIGS. 6 and 7.

Note that in FIGS. 6 and 7, the frequency of 2.5 MHz is used to providebackscatter modulation for logical “0” symbols 602 and 702, and in FIGS.8 and 9 the frequency of 3.75 MHz is used to provide backscattermodulation for logical “1” symbols 802 and 902. These frequencies areprovided by illustrative purposes, and the present invention isapplicable to the use of alternative frequencies for backscattermodulation.

Backscatter symbols 802 and 902 shown in FIGS. 8 and 9 each start atdistinct times. These distinct start times are attributable to thedistinct rising edge times (i.e., T_(A) and T_(B)) associated with thereader-originated “0” and “1” symbols described above with reference toFIGS. 3 and 4.

Accordingly, note that when the description below refers to thetransmission from tag 102 of “0” and “1” bits, signals, or symbolsduring binary traversals, these refer to the transmission of “0” and “1”backscatter symbols 602, 702, 802, and 902 as described above.Furthermore, when the description below refers to transmission fromreader network 104 of “0” and “1” bits, signals, or symbols duringbinary traversals, these refer to the transmission of “0” and “1”symbols 302 and 304 as described above.

In a preferred embodiment of the present invention, the reader signal“NULL” symbol 502 shown in FIG. 5 is not defined to have backscatterpresent in the S_(high) state due to tag 102. In alternativeembodiments, tag 102 may introduce backscatter in response to a “NULL”signal.

Reader network 104 determines the value of the bit or symbol that wasbackscatter modulated by tag 102. Reader network 104 samples a receivedsignal for backscatter modulation produced by one or more tags 102 ofthe population of tags 120. In a preferred embodiment, reader network104 samples the received signal at a timing point T_(BS) to determinewhether a backscatter modulated symbol was received. FIG. 26B shows therelative spacing of timing points T0, T1, T2, T_(A), T_(B), T_(C), andT_(S), to timing point T_(BS). As shown in FIG. 26B, timing point T_(BS)is preferably located between timing points T_(C) and T2. T_(BS) shouldbe located at a point after T_(C) in the received symbol wherebackscatter modulation has begun, and has time to propagate through thenecessary components of the receiver of reader network 104 to bedetected. T_(BS) should also be located at a point before T2 so that thereceived backscatter modulated symbol has not finished, and such thatthe length of received symbols can be as short as possible to increasethe read rate of tags 102.

1.2.3 Storage of Timing Points

Reader network 104 creates and coordinates timing points T0, T1, T2,T_(A), T_(B), T_(C), T_(S), and T_(BS). Timing points T_(A), T_(B),T_(C), T_(S), and T_(BS) are shown in FIG. 26B relative to timing pointsT0, T1, and T2. As described above, timing points T0, T1, T2, T_(A),T_(B), T_(C), T_(S), and T_(BS) relate to various timing characteristicsof the present invention, as described above. The timing points arestored so they may be used to maintain consistent timing duringcommunication between reader network 104 and tag 102. In an embodiment,reader network 104 stores the timing points, and conveys one or more ofthem to tags 102, as described elsewhere herein.

In a preferred embodiment, all of timing points T0, T1, T2, T_(A),T_(B), T_(C), T_(S), and T_(BS) are dynamic and adjustable by the readernetwork 104 and tags 102, subject to the requirements of the particularenvironment.

For example, in embodiments, a first reader in a reader network 104 mayuse timing characteristics different from those used by a second readerin the same or different reader network 104 operating in the samelocality to communicate with one or more of the same tags 102. Forexample, the first reader may lengthen the duration of one or more ofT_(A), T_(B), and T_(C) to give tag 102 more time to read symbols in anoisy environment. Conversely, the first reader may shorten the durationof one or more of T_(A), T_(B), and T_(C) to allow for the fasterreading of a large number of tags 102, relative to the second reader.After a binary traversal performed by the first reader network 104 iscomplete, the second reader may change the duration of one or moreT_(A), T_(B), and T_(C) before performing a binary traversal of tag 102.

Furthermore, according to embodiments, timing characteristics for aparticular tag 102 used for communication with a first reader network104 may be different from those used for communication with a secondreader network 104. For example, according to processes described below,the first reader network 104 may provide longer values of one or more ofT0, T1, and T2 to tag 102 to allow tag 102 to be able to read longersymbols in a noisy environment. Conversely, the first reader network mayprovide shorter values for one or more of T0, T1, and T2 so that tag 102is able to read shorter symbols, and hence can receive symbols morerapidly. After a binary traversal performed by the first reader network104 is complete, the second reader network 104 may provide differentvalues of one or more of T0, T1, and T2 to the tag 102 before performinga binary traversal of tag 102.

In another embodiment, once a binary traversal has begun, but has notcompleted, a reader network 104 may adjust one or more of timing pointsT_(A), T_(B), T_(C), T_(S), and T_(BS). Such an adjustment may beperformed as necessary to accommodate a noisy RF environment and otherconcerns that reader network 104 may have at the time. Reader network104 may also provide different values for one or more of timing pointsT0, T1, and T2 to tag 102 to reconfigure timing characteristics of tag102 in the midst of a binary traversal, as necessary.

2. Tag Embodiments According to the Present Invention

2.1 Structural Description of a Tag

2.1.1 Structural Overview

FIG. 10 is a block diagram of a tag 102, according to an embodiment ofthe present invention. Tag 102 includes an integrated circuit 1002, aplurality of pads 1004 a, 1004 b, 1004 c, and 1004 d, a capacitor 1006,an optional battery 1008, a first antenna 1010 a and a second antenna1010 b. These components are mounted or formed on a substrate 1001.These components are described in further detail below.

Pads 1004 provide electrical connections between integrated circuit 1002and other components related to tag 102. For instance, RF pad 1004 bestablishes a connection between integrated circuit 1002 and firstantenna 1010 a. Similarly, RF pad 1004 d provides a connection betweenintegrated circuit 1002 and second antenna 1010 b.

2.1.2 Capacitor/Battery

External power pad 1004 c and ground pad 1004 a establish connections toprovide integrated circuit 1002 with an operating voltage. As shown inFIG. 10, a capacitor 1006 is coupled between pads 1004 c and 1004 a.Capacitor 1006 stores operating voltage and power obtained through powerharvesting circuitry within integrated circuit 1002. This powerharvesting circuitry converts low-voltage oscillating RF energy thatintegrated circuit 1002 receives through antennas 1010 a and 1010 b intoa higher voltage direct current (DC) signal. Further details regardingsuch power harvesting techniques are provided below.

An optional battery 1008 or other power source may also be coupledbetween pads 1004 c and 1004 a. The use of battery 1008 makes thepresence of capacitor 1006 optional. In other words, capacitor 1006 mayeither be absent or coupled in parallel with battery 1008 (i.e., betweenpads 1004 c and 1004 a). When present, battery 1008 provides integratedcircuit 1002 with an operating voltage that is independent of theperformance of its power harvesting circuitry. Power harvestingcircuitry typically generates a DC voltage and current that is dependenton the level of available RF energy. Thus, as the physical distancebetween tag 102 and reader network 104 increases, the DC voltage levelthat is obtainable through power harvesting techniques decreases.

Accordingly, when integrated circuit 1002 relies solely on powerharvesting techniques for operational power, it may be possible for tag102 to receive information signals from reader network 104 that lackadequate energy to provide tag 102 with a sufficient operating voltage.However, such information signals may have a signal-to-noise ratio (SNR)that would be large enough for decoding if integrated circuit 1002 wereoperational. When employed, battery 1008 provides such an operationalvoltage. Therefore, the use of battery 1008 enables tag 102 tocommunicate with reader network 104 at greater distances, and/or inchallenging RF environments. Battery 1008 may be of a variety of types,both in chemical composition and form factor, including types that canbe printed directly on tag substrate 1001. A less expensivedischarge-only type of battery will have a certain useful life beforebecoming unable to supply enough operating power to tag 102.Alternatively, a small rechargeable battery may support the operation oftag 102 while in challenging RF environments. The rechargeable batterycould also be recharged in an RF environment sufficient to drive thepower harvesting function of tag 102. In embodiments, alternativesources for harvesting energy from the environment include, but are notlimited to solar cells, piezoelectric materials that convert vibrationto voltage, and other sources known to persons skilled in the relevantarts.

In an alternative embodiment, as illustrated in FIG. 11, tag 102 mayinclude components used to receive information from at least one sensor1111. In an embodiment, an analog to digital converter (A/D) 1180receives an analog sensor signal from sensor 1111, and converts theanalog sensor signal to digital. Sensor 1111 may be internal or externalto integrated circuit 1002. If sensor 1111 is external to integratedcircuit 1002, sensor 1111 will couple to a connection pad 1104 d, whichis coupled to A/D 1180, as shown in FIG. 11. Power bus 1054 providespower to A/D 1080. As shown in FIG. 11, a single RF pad, RF1 pad 1004 b,is present for coupling with antenna 1010 a, and a single connection pad1104 d is present for coupling with sensor 1111. In further embodiments,one or more additional connection pads may be present to couple withsensor 1111, and/or with additional sensors. Additional antenna pads mayalso be present. Furthermore, when present battery 1008 or capacitor1006 may be internal to integrated circuit 1002, and therefore, groundpad 1004 a and power pad 1004 c may not be present. When sensor 1111 isinternal to integrated circuit 1002, sensory pad 1104 d may not bepresent. The present invention is applicable to any combination of theseantenna and sensor configurations.

2.1.3 Orientation Insensitivity

The strength of RF signals received from reader network 104 is dependentupon the design of the antenna that collects the energy from the RFenvironment. Current antenna design theory and practice show that energyreceived is in part a function of the gain and the width of reception(beam width). Gain is inversely proportional to beam width in that asantennas are adjusted to receive from broader directions, they receiveat lesser gain, or lesser power as a result. Conversely, as antennas areadjusted for receipt of maximum power or gain, the power will only beattainable at a very specific orientation with respect to thetransmitting antenna. This may result in an orientation sensitivity fora tag antenna with respect to a reader antenna that can greatly reducethe operational distance for a non-optimum orientation.

This fundamental problem has long existed with respect to RFIDtechnology, and a solution has been desired. Resultingly, an industrygoal is to determine how to maintain a maximum read distance (which isdirectly related to antenna gain) while minimizing or removingaltogether sensitivity to tag antenna orientation (i.e., the directionof the tag antenna with respect to the reader antenna). Currentlyavailable devices exhibit reasonable read ranges, but only in certaintag orientations. This is not desirable to the majority of marketslooking for RFID products.

An advantage of the present invention is the ability of integratedcircuit 1002 to handle multiple antenna inputs. For example, in thepreferred embodiment shown in FIG. 10, first and second antenna pads1004 b and 1004 d are present to accommodate first and second antennas1010 a and 1010 b, respectively. This allows multiple standarddirectional antennas to be oriented on substrate 1001 such that theaverage gain over all orientations is increased with respect to eachantenna separately. In a preferred embodiment, antenna 1010 a isoriented such that its maximum gain is in a direction that correlateswith the minimum gain of antenna 1010 b. For example, when usingstandard dipole designs, antenna 1010 a would be oriented at a 90 degreeangle with respect to antenna 1010 b on the same substrate (Z axisremains constant).

Another such use of multiple antenna inputs would be to simplify wideband receiving antennas. Again, a similar problem exists in that thewider frequency agility of a given antenna design will reduce the gain,or collected energy of a single antenna. Allowing multiple antennas tobe designed each for their own distinct frequency bands allows tag 102to function in the overall wide band with more power in each of thedistinct frequencies than a more complicated single wide band antennadesign would allow. It is desired that RFID products are capable ofoperating worldwide. Distinctly different frequency bands may be presentin each country to operate in a license free environment. Hence, theability to use multiple antenna designs for multiple frequency bands isan advantage of the present invention.

2.1.4 Tag Substrate

Integrated circuit 1002 may be implemented across more than oneintegrated circuit chip, but is preferably implemented in a single chip.The one or more chips of integrated circuit 1002 are created in one ormore wafers made by a wafer fabrication process. Wafer fabricationprocess variations may cause performance differences between chips. Forexample, the process of matching inductances of a chip may be affectedby fabrication process differences from wafer-to-wafer, lot-to-lot anddie-to-die.

Integrated circuit 1002 is mounted to substrate 1001. First and secondantennas 1010 a and 1010 b are printed on substrate 1001. In anembodiment, the materials used for substrate 1001 are 3–5 Mil MYLAR®- orMYLAR®-like materials. The MYLA® related materials are preferably usedbecause of their relatively low dielectric properties, as well as theirbeneficial printing properties. Conductive inks used to print an antennadesign are cured at very high temperatures. These high temperatures cancause standard polymers to degrade quickly as well as become veryunstable to work with.

An antenna design is printed on substrate 1001 with the conductive inks.In an embodiment, the conductive inks are primarily silver particlesmixed with various binders and solvents. For example, binders andsolvents currently manufactured by DuPont Corporation may be used. Theconductive inks can have different silver particle loads, which allowscreation of the desired level of conductivity. Once an antenna isprinted, the resistance or “Q” may be determined from the antennadesign. A matching circuit may then be determined that allows a match ofthe surface of antennas 1010 a and 1010 b to first and second antennapads 1004 b and 1004 d, respectively, providing an effective read rangefor tag 102. In alternative embodiments, antenna substrates of any typeor manufacture may be used. For example, subtractive processes thatobtain an antenna pattern by etching, or by removing material from acoated or deposited substrate may be used. In a further alternativeembodiment, the antenna substrate may be eliminated altogether, and theantenna(s) may be incorporated directly into the integrated circuit.

Note that conductive materials by their own nature tend to oxidize,resulting in an oxide material forming on a surface of the conductivematerial. The oxide material can be conductive or non-conductive.Non-conductive oxides are detrimental to RF (UHF) performance, as theycan significantly cause an antenna to detune. Therefore, in a preferredembodiment, a conductive material may be chosen that tends to oxidizewith a conductive oxide. For example, the conductive material may besilver, nickel, gold, platinum, or other Nobel metal, as opposed tocopper or aluminum, which tend to oxidize in a non-conductive fashion.However, in alternative embodiments, any suitable material may be usedfor the conductive ink, including conductive materials that tend tooxide in a non-conductive fashion, such as those listed above.

2.1.5 Integrated Circuit

As shown in FIG. 10, integrated circuit 1002 includes a data programmingunit 1020, a state machine 1024, a timing subsystem 1023, and an RFinterface portion 1021. In an embodiment, data programming unit 1020permanently stores information, such as a tag identification number aswell as other data. Alternatively, the information may be storedtemporarily. The storage of information in data programming unit 1020may be performed using a variety of techniques. For example, many typesof laser programming techniques are available and may be used. Focusedion beam (FIB) techniques are also available and applicable to thepresent invention. Each of these exemplary techniques typically is usedduring or soon after production of integrated circuit 1002. In anembodiment, redundant structures for storing bits of information usingthe laser programming techniques can be used to reduce the effect ofsingle cell programming process errors. Similarly, in anotherembodiment, dual cells of a programming bit can be implemented in such afashion that would require a cell to be programmed with all cases of ‘0’or ‘1’ bits to allow for a reduced power detection circuitry and/or anend of variable ID length detection. Other techniques for the permanentstorage of an identification number in integrated circuit 1002 are alsowithin the scope of the present invention.

State machine 1024 may include logic, a processor, and/or othercomponents that controls the operation of RFID tag 102. In anembodiment, state machine 1024 is implemented with digital circuitry,such as logic gates. Further details regarding state machine 1024 areprovided below with reference to FIG. 12A.

RF interface portion 1021 is coupled to first and second antennas 1010 aand 110 b to provide a bidirectional communications interface withreader network 104. In an embodiment, RF interface portion 1021 includescomponents that modulate digital information symbols into RF signals,and demodulate RF signals into digital information symbols. Furthermore,RF interface portion 1021 includes components that convert a wide rangeof RF power and voltage levels in the signals received from first andsecond antennas 1010 a and 1010 b into usable signals. For example, thesignals may be converted to the form of transistor usable direct current(DC) voltage signals that may have substantially higher or lower levelsthan output by first and second antennas 1010 a and 1010 b.

FIG. 10 shows that RF interface portion 1021 features two sets of thesame components. RF interface portion 1021 includes a first and a secondreceiver 1030 a and 1030 b, a first and a second charge pump 1032 a and1032 b, and a first and a second modulator 1034 a and 1034 b. Each ofthese components is coupled to a respective one of first and secondantennas 1010 a and 1010 b. First receiver 1030 a, first charge pump1032 a, and first modulator 1034 a are each coupled to first antenna1010 a. Second receiver 1030 b, second charge pump 1032 b, and secondmodulator 1034 b are each coupled to second antenna 1010 b.

First and second charge pumps 1032 a and 1032 b operate to provideintegrated circuit 1002 with an operational voltage. As shown in FIG.10, first charge pump 1032 a receives first RF signal 1050 a from firstantenna 1010 a. First charge pump 1032 a converts first RF signal 1050 ainto a first DC voltage signal 1052 a. Similarly, second charge pump1032 b receives second RF signal 1050 b from second antenna 1010 b andproduces a second DC voltage 1052 b. First and second DC voltage signals1052 a and 1052 b are combined at a node 1053. Node 1053 produces anoperational voltage signal/power bus 1054, which provides power tointegrated circuit 1002. Although FIG. 10 shows operational voltagesignal 1054 only being sent to state machine 1024, power bus 1054 ispreferably a bus that provides power to one or more of the othercomponents within integrated circuit 1002 as required.

Further details regarding implementations of first and second receivers1030 a and 1030 b and first and second charge pumps 1032 a and 1032 bare provided below.

First and second modulators 1034 a and 1034 b are coupled to first andsecond antennas 1010 a and 1010 b, respectively. In an embodiment, eachof first and second modulators 1034 a and 1034 b includes a switch, suchas a single pole, single throw (SPST) switch. The switch changes thereturn loss of the respective one of first and second antennas 1010 aand 1010 b. The return loss may be changed in a number of ways. Forexample, when the switch is in its ‘on’ condition, the RF voltage at therespective antenna may be set lower than the RF voltage at the antennawhen the switch is in its ‘off’ condition by predetermined percentage(e.g., 30 percent). This may be accomplished by a variety of methodsknown to persons skilled in the relevant arts.

Each of first and second modulators 1034 a and 1034 b may drive itscorresponding switch at the frequency of clock signal 1064 or at thefrequency of clock signal 1066. Modulation with either of these clocksignals creates upper and lower side bands in the energy that isreflected by the respective antenna. Thus, when receiving a signal fromreader network 104, tag 102 backscatters energy in frequencies that arenot transmitted by reader network 104. This feature enables the firstfrequency to designate a logical “1” bit and the second frequency todesignate a logical “0” bit. Integrated circuit 1002 includes afrequency selector 1040. Frequency selector 1040 outputs two or morepossible frequencies on a frequency signal 1040. First and secondmodulators 1034 a and 1034 b receive frequency signal 1040. Frequencysignal 1040 determines at which frequency first and second modulators1034 a and 1034 b operate.

As shown in FIG. 10, two sets of modulator, charge pump, and receivercomponents are present in RF interface portion 1021: first and secondmodulators 1024 a and 1024 b, first and second charge pumps 1032 a and1032 b, and first and second receivers 1030 a and 1030 b. Note that thepresent invention is applicable to any number of one or more sets ofthese components, and related antennas. Accordingly, the presentinvention allows for a single RF signal to be received and processed,and for any number of two or more RF signals to be simultaneouslyreceived and processed. The ability to receive multiple RF input signalsfacilitates a unique method of the present invention that allows forinsensitivity to the orientation of a responding tag 102, as furtherdescribed elsewhere herein.

As shown in FIG. 10, first and second charge pumps 1032 a and 1032 boutput electricity onto power bus 1054. In an embodiment, when power bus1054 receives two DC voltages at node 1053, the DC voltages do notconflict. Instead, the higher voltage of the two DC voltages dominates,and supplies more power to capacitor 1006 and other components requiringpower in integrated circuit 1002.

First receiver 1030 a outputs a first received signal 1056 a to statemachine 1024, and second receiver 1030 outputs a second received signal1056 b to state machine 1024. In such an embodiment where RF interfaceportion 1021 includes two sets of components, a logical ‘OR’ing functionmay be applied to first and second received signals 1056 a and 1056 b instate machine 1024. As a result, only one of first and second receivers1030 a and 1030 b is required to output an edge on first and secondreceived signals 1056 a and 1056 b to indicate that data has beenreceived. The detection of two or more edges reinforces the duplicatereceived information. In an embodiment, state machine 1024 processes thefirst of first and second received signals 1056 a and 1056 b thatprovides an edge. Hence, in the present invention, multiplesimultaneously received signals are logically ‘OR’ed into a singlesignal, with the first signal being considered dominant.

State machine 1020 accesses data processing unit 1020 over dataprocessing unit bus 1076 to determine whether a logical “1” or “0” is tobe transmitted by tag 102. More specifically, state machine 1020accesses one or more bits of the identification number stored in dataprocessing unit 1020. The one or more accessed bits allow state machine1020 to determine whether reader network 104 is addressing thisparticular tag 102 during the present portion of the current binarytraversal, and what response, if any, is appropriate. Accordingly, statemachine 1024 outputs a frequency selection signal on first and secondcontrol signals 1060 a and 1060 b. The frequency selection signalindicates which of a “0,” a “1,” or other backscatter symbol is to betransmitted from tag 102. First and second modulators 1034 a and 1034 breceive first and second control signals 1060 a and 1060 b,respectively. In the embodiment of FIG. 10, first and second controlsignals 1060 a and 1060 b direct first and second modulators 1034 a and1034 b to perform one of at least the following three actions: (1)perform backscatter modulation using the frequency of clock signal 1064,(2) perform backscatter modulation using the frequency of clock signal1066, or (3) do nothing. For (1) and (2), first and second modulators1034 a and 1034 b preferably perform modulation in tandem at theselected frequency. Hence, the frequency selection signal of first andsecond control signals 1034 a and 1034 b may be the same physicalsignal. Accordingly, in a preferred embodiment, first and secondantennas 1010 a and 1010 b perform backscatter at the same frequency.

In a two-antenna embodiment for tag 102, one of first and secondantennas 1010 a and 1010 b may be positioned in a better orientation forpower than the other antenna, relative to reader network 104. Thisantenna will typically provide more backscatter energy for the antennaof reader network 104 to detect. Hence, the better oriented antenna oftag 102 will typically transmit signals that prevail over signalstransmitted from the other antenna of tag 102. Note that this principleis also applicable to greater numbers of antennas for tag 102 than justtwo.

As shown in FIG. 10, timing subsystem 1023 includes an oscillator 1026,a successive approximation register (SAR) 1022, a counter 1028, a firstdivider 1036, and a second divider 1038. Oscillator 1026 generates amaster clock signal 1062 having a master clock frequency, such as 7.5MHz. Master clock signal 1062 is received by first divider 1036 and bysecond divider 1038. First and second dividers 1036 and 1038 each dividethe frequency of master clock signal 1062, and output first and secondclock signals 1066 and 1064, respectively.

First and second clock signals 1066 and 1064 each have a frequency thatis less than the frequency of master clock signal 1062. For instance,first divider 1036 may divide the frequency of master clock signal 1062by a factor of two. Hence, the frequency of second clock signal 1064 isone-half of the frequency of master clock signal 1062. Second divider1038 may divide the frequency of master clock signal 1062 by a factor ofthree. Hence, the frequency of first clock signal 1066 is one-third ofthe frequency of master clock signal 1062. Accordingly, when thefrequency of master clock signal 1062 is 7.5 MHz, the frequency ofsecond clock signal 1064 is 3.75 MHz and the frequency of first clocksignal 1066 is 2.5 MHz.

Clock signals 1064 and 1066 are received by various components ofintegrated circuit 1002. In an embodiment, first clock signal 1066 isused as the system clock signal for integrated circuit 1002. First clocksignal 1066 is lower in frequency, and therefore promotes lower powerusage by components of tag 102. In an embodiment, first clock signal1066 is received by counter 1028. Counter 1028 increments an internalregister at a rate that corresponds to the frequency of first clocksignal 1066, to generate a count value. FIG. 10 shows counter 1028 as anine bit binary counter. However, counter 1028 may have different bitwidths and configurations as dictated by the particular application.

The count value of counter 1028 may be cleared upon the occurrence ofcertain conditions. For example, counter 1028 may be cleared during datacalibration procedures. Data calibration procedures are described ingreater detail below with reference to FIGS. 26–28D.

Oscillator 1026 is coupled to SAR 1022 by a control interface 1070.Successive approximation register 1022 sends a control signal tooscillator 1026 across interface 1070 to adjust (i.e., to calibrate) thefrequency of master clock signal 1062, and hence to adjust the frequencyof first clock signal 1066. FIG. 10 shows control interface 1070 havingeight parallel control signals. However, any number of one or morecontrol signals may be used for interface 1070. The operation of SAR1022 and oscillator 1026 is described in greater detail below withreference to FIGS. 18–25C.

2.2 Functional Description of a Tag

2.2.1 Operational States of a Tag

Tag 102 can exist in various operating states. Each of these operatingstates describes a mode of operation for tag 102. Upon the occurrence ofcertain events, tag 102 can transition from one operating state toanother. For example, upon occurrence of an event, tag 102 cantransition from a present operating state, which is the operating statethat tag 102 is operating in when the event occurs, to a new operatingstate, as dictated by the combination of the present operating state andthe event. In an embodiment, these events can be classified in twocategories: Data events and time-based events. Data events are triggeredby the detection of edges from transmissions of reader network 104, suchas the transition from S_(low) to S_(high) and vice versa. Time-basedevents are derived from a passage of a certain period of time, such asmay be indicated by a counter overflow. In a preferred embodiment, atimer or counter is reset (e.g., the timer or counter outputs a zerocount) upon detection of a data event. Time-based events may beconsidered to be indications that no data events have occurred over aparticular period of time.

In FIGS. 12B, 12C, and 12D, possible combinations of time-based and databased events are shown. In FIG. 12B, a data transition from S_(low) toS_(high) resets the counter or timer to zero at time T_(CS). At the endof a period of time indicated by Tov, where the counter overflows totrigger an elapsed time event, the event is considered a master resetevent 1220. Master reset event 1220 occurs on a timer or counteroverflow when the data is in an S_(high) state. In FIG. 12C, a datatransition from S_(high) to S_(low) also resets the timer to zero attime T_(CS). At the end of the period of time indicated by Tov, atime-based event occurs, which is considered a master dormant event1221. This occurs because the data value has remained in the S_(low)state, as opposed to transitioning to the S_(high) state, as shown inFIG. 12B. In an embodiment, this event is applicable to battery poweredtags. Power drawn from the battery of battery powered tags may bereduced after the master dormant event 1221 occurs as shown in FIG. 12C.

FIG. 12D shows a preferred embodiment of the present invention thatallows for the conservation of battery power for tag 102 when in aninactive mode due to the input data remaining in the S_(high) state. Forexample, a constant level of ambient noise on a received signal mayappear to be an S_(high) state in certain situations, and thereforecould activate tag 102. In another example, a reader network 104 mayinadvertently enter a state where it is outputting an RF transmissionwith no modulation (e.g., a constant wave (CW) emission). FIG. 12D showsa force low event 1250 that will force the state of the data line to beconsidered as S_(low). In a preferred embodiment, data events will besuppressed while the data line is being forced low after force low event1250. However, the counter or timer will be reset to allow the eventshown in FIG. 12C to be generated. If no additional edges are detectedin the signal received from reader network 104, then the condition shownin FIG. 12C will generate a master dormant event 1221, thereby placingtag 102 into a power conservation mode.

It is also important to note the time length of the time period Tov,which represents the length of time for an overflow of counter 1028. Asa tag 102 initially powers up, and the frequency of the oscillatordriving the timer function is not calibrated, the actual time period ofTov may vary between tags by +−50%. This variation is due to variationsin fabrication processes and due to ambient environmental conditionssuch as temperature. In a preferred embodiment, Tov is ideally equal to400 μS. Under real operating conditions, this value for Tov may varybetween 200 and 600 μS. This variation does not include the timenecessary for tag 102 to power up and begin counting.

Receipt of a master reset event 1220 (i.e., received signal of lengthgreater than Tov) may be used to cause a tag 102 to enter a calibrationmode, for example. In an embodiment, the length Tov for a particular tag102 is multiplied four times after the receipt of the first master resetevent 1220. This adjustment of Tov by tag 102 enables reader network 104to initiate a new calibration procedure at any time with a new tag 102that enter its communications range without affecting existing tags 102.For example, a new calibration procedure is preferably initiated afterreader network 104 has already interrogated all tags 102 within itscommunications range. Thus, when new tags 102 enter the communicationrange of reader network 104, reader network 104 may re-transmit theshorter type of master reset signal. This re-transmission of the shortermaster reset signal initiates calibration procedures and subsequentprotocol exchange with the new tags 102. The existing tags 102 do notre-enter calibration mode, because they now require the longer masterreset signal to enter calibration mode.

FIG. 12A illustrates various operating states in a state diagram for tag102, according to an embodiment of the present invention. In FIG. 12A,each operating state is shown as an oval, and transitions betweenoperating states are shown as connections between the ovals. Thetransitions are annotated with text that describes a correspondingevent. Located at the bottom of FIG. 12A are two disjoint statetransitions that are indicative of the interrupting time-based masterreset and master dormant events. Note that the two disjoint statetransitions are not shown integrated into the state diagram to aid thereadability of the state diagram. The two disjoint state transitions aretransition options that are available at each state, to transition fromany state to the final target state.

The paragraphs below describe the operating states and the respectivetransitions shown in FIG. 12A. These particular states and transitionsare presented by way of example only. Additional and alternativeoperating states, transitions, and transition causing events can beemployed without departing from the spirit and scope of the presentinvention.

The first state is a dormant state 1202. During dormant state 1202, tag102 is largely inactive. Therefore, power is conserved during dormantstate 1202. Tag 102 enters dormant state 1202 upon powering up, afterreceipt of a master dormant event, and at other times described below.When tag 102 is in dormant state 1202, first and second receivers 1030 aand 1030 b and first and second charge pumps 1032 a and 1032 b arecoupled to first and second antennas 1010 a and 1010 b, respectively, toreceive energy and data from reader network 104.

For example, while in dormant state 1202, first and second charge pumps1032 a and 1032 b supply power that is used to charge capacitor 1006.The power is generated from RF transmissions received by first andsecond antennas 1010 a and 1010 b. The RF transmissions may originatefrom reader network 104 while it is performing interrogation operationsunrelated to tag 102. The RF transmissions may also originate from othersources of RF energy. The charging of the capacitor 1006 enables tag 102to achieve an operating voltage. When this operating voltage is reached,tag 102 has the capability to function in the manner described herein.

As shown in FIG. 12A, tag 102 transitions from dormant state 1202 into acalibration state 1204 upon the master reset event described in FIG.12B. Additionally, tag 102 may transition from other states tocalibration state 1204. This transition is shown in FIG. 12A as masterreset event 1220. In an embodiment, dormant state 1202 is only able totransition to calibration state 1204. No other data events will resultin a transition from dormant state 1202. In alternative embodiments,events may cause transitions from dormant state 1202.

In calibration state 1204, tag 102 initializes its timing circuitry. Inan embodiment, in calibration state 1204, tag 102 will not generate dataevents “0,” “1,” and “NULL,” as they have not yet been defined. Instead,in calibration state 1204, tag 102 performs an oscillator calibrationprocedure and a data calibration procedure. The oscillator calibrationprocedure involves tag 102 receiving multiple oscillator calibrationpulses from reader network 104, defined herein as edge transition (data)events. Specific timing is provided between the edge transition events.Similarly, the data calibration procedure involves tag 102 receivingmultiple data calibration pulses from reader network 104. The datacalibration pulses are also defined as edge transition events withspecific timing. Example data calibration and oscillator calibrationtechniques are described in further detail below.

Before tag 102 completes the oscillator calibration procedure, thesystem timer or counter operates at an uncalibrated rate. As describedabove, the uncalibrated rate may be within a +/−50% tolerance of acalibrated system timer rate. This variation may be in part due toprocess variations of standard integrated circuit manufacturing and toambient environmental conditions such as temperature. Accordingly, anoverflow period used to designate master reset signals is within apredetermined tolerance. For example, in an embodiment, reader network104 provides a reset signal causing a master reset event 1220 that is ofa duration of time 50% greater than a center time duration. In apreferred embodiment, the center time duration may be 400 μS, so thatthe 50% greater time duration that occurs due to master reset event 1220is 600 μS. Hence, reader network 104 ensures that tags 102 recognize amaster reset signal 1220, regardless of their process variations,ambient temperature, and oscillator tolerances.

As shown in FIG. 12A, tag 102 may transition from calibration state 1204to dormant state 1202 upon the occurrence of event 1222. In anembodiment, event 1222 is defined by the reception of a signal that arenot representative of timing signals expected by tag 102. For instance,in a preferred embodiment, oscillator calibration signals are defined as8 pulses of equal length. If the oscillator calibration pulses receivedby tag 102 are significantly unequal or not within an expected range oflengths, the pulses may be considered invalid, causing occurrence of anevent 1222. Hence, when tag 102 receives signals that do not causesuccessful oscillator calibration or data calibration procedures, thiscauses an event 1222 to occur.

After successful completion of the oscillator calibration procedure,which results in a tuned oscillator, and the data calibration procedure,which results in defined data symbols, tag 102 transitions fromcalibration state 1204 to a command state 1206. This transition is shownin FIG. 12A as transition or event 1224. After data calibration, tag 102expects to receive defined data symbols from reader network 104. Thedata symbols are defined as data “0,” data “1,” and data “NULL.” Masterreset and master dormant events may occur at any time, and areimmediately processed after occurring.

During command state 1206, tag 102 expects a command from reader network104 in the form of a data symbol. Such a command directs tag 102 toenter either a tree traversal state 1208 or a superposition state 1210.In a preferred embodiment, the command is a single bit. For example,receipt of a logical “0” symbol 302 from reader network 104 may directtag 102 to enter tree traversal state 1208. However, receipt of alogical “1” symbol 402 from reader network 104 may direct tag 102 toenter superposition state 1210. The transition from command state 1206to tree traversal state 1208 is shown in FIG. 12A as event 1230, whilethe transition from command state 1206 to superposition state 1210 isshown as event 1232. In an embodiment, the receipt of a logic “NULL”symbol 502, as shown in FIG. 5, does not effect the state of tag 102 incommand mode. This is shown as event 1226 in FIG. 12A.

When operating in tree traversal state 1208, tag 102 transmits itsidentification number to reader network 104 according to a binarytraversal protocol that enables reader network 104 to quicklyinterrogate a population of tags 120. The binary traversal protocol isdescribed in greater detail below with reference to FIGS. 13–16.

Tag 102 may enter a mute state 1212 from tree traversal state 1208. Thisis shown in FIG. 12A as transition 1238. In mute state 1212, tag 102receives data from reader network 104. However, when in mute state 1212,tag 102 will provide no response until a data “NULL” signal is receivedby tag 102. The data “NULL” signal returns tag 102 to command state 1206via event 1244. The mute state 1212 disables a tag 102 from respondingto a particular request for an ID from a population of tags 120. In apreferred embodiment, reader network 104 does not directly address a tag102, or population of tags 120, in an effort to disable them fromcommunicating to reader network 104. Tag 102 determines whether it isacceptable to continue transmissions to reader network 104, and when itis necessary to enter mute state 1212. Information necessary todetermine these actions is provided implicitly by reader network 104 totag 102, as described herein.

Alternatively, tag 102 may return to dormant state 1202 from treetraversal state 1208. Tag 102 transitions from tree traversal state 1208to dormant state 1202 upon receipt of a data “NULL” symbol 502 fromreader network 104. In a preferred embodiment, receipt of the data“NULL” symbol 502 occurs after reader network 104 has determined thatall desired information has been obtained from tag 102, and all othertags 102 of the tag population have transitioned into a state ofinactivity (i.e., either dormant state 1202 or mute state 1212).Optionally, in an embodiment, tag 102 may also transition itself fromtree traversal state 1208 to dormant state 1202 when tag 102 hascompleted transmission of its identification number to reader network104. In another embodiment, tag 102 may transition to a deep dormantstate at this point such that the amount of power used by tag 102 is ata minimal level necessary to maintain operation in the state. Hence,transmitted RF energy is allowed to pass by or through a tag 102 in thedeep dormant state to other tags 102 in close proximity so that they canobtain more power for operation.

Note that during tree traversal operations, one or more tags 102 mayfind themselves active, and in state 1208, or temporarily inactive, andin state 1212. One or more other tags 102 that have been processed willbe in dormant state 1202. Reader network 104 may then collectivelyaddress the full population of tags 120 by sending a data “NULL” symbol502. Tags 102 that were active in state 1208 will then transition to thedormant state 1202 via event 1242, joining one or more tags 102 indormant state 1202. However, one or more tags 102 that were temporarilyinactive in mute state 1212 will transition back to activeparticipation, in command state 1206. The transmission of a single logicsymbol from the reader network 104 to the tag population 102 causes allof these actions. Accordingly, this is an example of an implicitinstruction from reader network 104.

When operating in superposition state 1210 shown in FIG. 12A, tag 102receives information from reader network 104. Tag 102 responds to readernetwork 104 when designated portion(s) of its identification match theinformation received from reader network 104. Superposition state 1210allows acquisition of information regarding the entire population oftags 120 that is within communications range of reader network 104. In apreferred embodiment, this information is used by reader network 104 toquickly ascertain the most efficient tag interrogation algorithm to usefor the particular tag environment. With respect to a tag 102, thedifferences between tree traversal state 1208 and superposition state1210 are at least two-fold. First, information received from readernetwork 104 and information to be transmitted by tag 102 is compared todetermine whether to transmit the information. Second, if theinformation does not match, tag 102 does not enter into mute state 1212,but just ‘skips’ this one particular piece of information.

The receipt by tag 102 of a data “NULL” symbol 502 from reader network104 affects the operation of tag 102. A data “NULL” symbol 502 isdefined according to the particular operating state in which tag 102 isoperating. In particular, tag 102 recognizes the data “NULL” symbol 502when it is operating in one of command state 1206, tree traversal state1208, superposition state 1210, and mute state 1212. An exemplary data“NULL” symbol 502 is described above with reference to FIG. 5.

Tag 102 may transition between various operating states upon the receiptof a data “NULL” signal. For instance, when tag 102 is operating insuperposition state 1210, receipt of a data “NULL” symbol 502 causes tag102 to transition to command mode 1206. This transition is shown in FIG.12A as event 1228. However, when tag 102 is operating in tree traversalmode 1208, a data “NULL” causes tag 102 to transition to dormant state1202. FIG. 12A illustrates this transition as event 1242. When tag 102is operating in mute state 1212, receipt of a data “NULL” causes tag 102to transition to command state 1206. This transition is shown in FIG.12A as event 1244. Hence, although reader network 104 issues a data“NULL” symbol 502, it is the responsibility of the population of tags120 to interpret this symbol and act appropriately according to thecurrent state of each tag 102. Accordingly, this is another example ofthe implicit command set issued by reader network 104, according to apreferred embodiment of the present invention.

3. Communications Protocols According to the Present Invention

3.1 Binary Traversal Protocol

When operating in tree traversal state 1208, tag 102 communicates withreader network 104 according to a binary traversal protocol. Thisprotocol enables reader network 104 to rapidly retrieve information thatis associated with every tag 102, such as an identification number,within its communications range.

In the description below, reader network 104 transmits logical symbolsto tag 102 from the symbol set of logical “0” symbol 302, logical “1”symbol 402, and “NULL” symbol 502, which are respectively describedabove with respect to FIGS. 3–5. Furthermore, in the description below,tag 102 is described as responding to reader network 104 usingbackscatter symbols. The backscatter symbols are included in thebackscatter symbol set of “0” backscatter symbol 602, “0” backscattersymbol 702, “1” backscatter symbol 802, and “1” backscatter symbol 902,which are respectively described above with respect to FIGS. 6–9. In thetext below, note that the particular backscatter symbol used by tag 102to respond depends on the symbol received from reader network 104, andis chosen from this set of backscatter symbols, as described above.

FIG. 13 provides a flowchart illustrating an example operation of thebinary traversal protocol from the perspective of a single tag 102,according to an embodiment of the present invention. By operatingaccording to the flowchart of FIG. 13, tag 102 responds to signals fromreader network 104 with a reduced level of required processing. Hence,tags 102 require fewer transistors, thereby consuming less power andoccupying less space, which reduces integrated circuit costs.

As shown in FIG. 13, the flowchart begins with step 1302. In step 1302,tag 102 is in dormant state 1202.

In step 1304, tag 102 receives a master reset signal from reader network104. Upon receipt of this signal, tag 102 moves from dormant state 1202to calibration state 1204, and operational flow proceeds to step 1306.

In step 1306, tag 102 is synchronized with reader network 104.Accordingly, in step 1306, tag 102 performs oscillator calibration withreader network 104, and performs data calibration with reader network104. These procedures are further described below.

In step 1307, tag 102 enters command state 1206 and operation proceedsto a step 1308.

In step 1308, tag 102 initializes the data to be transmitted. The datais retrieved from data programming unit 1020 shown in FIG. 10. Tag 102prepares to send the first bit.

In step 1309, tag 102 receives a command from reader network 104 toenter tree traversal state 1208. In a preferred embodiment, the commandis a single bit, such as a logical “0” symbol 302.

In step 1310, tag 102 sends the designated identification number bit toreader network 104. Accordingly, step 1310 may include tag 102 sending abackscatter modulated symbol to reader network 104, such as one of thesymbols described above with reference to FIGS. 6, 7, 8, and 9. Asdescribed below with reference to FIGS. 14 and 15, reader network 104receives the identification bit and determines which tag 102 (or tags102) to address next. This determination involves reader network 104transmitting a bit value (i.e., either a “0” or a “1”) that it considersvalid. With reference to tag 102 shown in FIG. 10, step 1310 may includestate machine 1024 referencing a least significant (LSB) tag bit pointerto the designated identification number bit stored in data programmingunit 1020 in FIG. 10.

In step 1312, tag 102 receives a next bit from reader network 104.

In step 1316, the tag determines if the bit received from the readernetwork 104 in step 1312 is a data “NULL” symbol 502. If the bit is adata “NULL” symbol 502, reader network 104 instructs tag 102 todiscontinue operations until the next reset, and operation passes tostep 1302. However, if the bit is not a data “NULL” symbol 502,operation passes to step 1318.

In step 1318, tag 102 compares the bit sent in step 1310 and the bitreceived from reader network 104 during prior step 1312. If the bits donot match, operation passes to step 1319. This is an example of theimplicit nature of the present invention. Transmitting a single datasymbol from reader network 104 causes each tag 102 in the population oftags 120 to contextually switch to one of several possible states. Thisimplicit operation contrasts with conventional “explicit” protocols. Inexplicit protocols, a command from a reader specifically addresses apopulation or subset population of tags. Hence, only the addressedpopulation or subset population of tags moves to a directed state.Subsequent commands are required to address the remaining tags. Thus,multiple operations are required by the reader to accomplish what theimplicit approach of the present invention can accomplish in a singleoperation.

In step 1319, tag 102 enters mute state 1212, where backscattertransmissions are suspended. However, tag 102 keeps track of datasymbols being sent by reader network 104.

In step 1320, tag 102 receives a data symbol from reader network 104.

In step 1321, after receipt of a symbol from reader network 104 in step1320, tag 102 determines if the symbol is a data “NULL” symbol 502. Ifit is not, operation passes to state 1319.

Upon receipt of a data “NULL” signal in step 1321, operation passes tostep 1307, where tag 102 transitions to command state 1206.

If the bit received from reader network 104 during step 1312 matches thebit sent by tag 102 during the prior step 1310, tag 102 remains in treetraversal state 1208. Operation then may proceed to an optional step1322, when present. If step 1322 is not present, operation proceeds tostep 1324.

In optional step 1322, tag 102 determines whether it has sent alldesired information (e.g., identification information) to reader network104. If all desired information has been sent, tag 102 has beencompletely read (i.e., interrogated), and operation passes to step 1302.If tag 102 determines in step 1322 that all desired information has notbeen sent to reader network 104, operation proceeds to step 1324.

In step 1324, tag 102 designates a next bit of its identification numberto be sent to reader network 104. For example, the designated next bitmay be the next significant bit to the previously selected bit (i.e.,either the bit next to the bit initially designated in step 1308, or thebit selected most recently in step 1324). Thus, step 1324 may includestate machine 1024 incrementing its tag bit pointer to the next mostsignificant bit position in data programming unit 1020 shown in FIG. 10.

After completion of step 1324, operation passes to step 1310. In step1310, tag 102 transmits the bit designated in step 1324 to readernetwork 104 as a backscatter symbol.

During traversal sequences, such as the traversal sequence describedabove with reference to FIG. 13, reader network 104 may employ varioustraversal termination techniques that each cause tag 102 to enterdormant state 1202. That is, such termination techniques will causeoperation of tag 102 to return to step 1302.

The ability to cause tag 102 to enter dormant state 1202 when desiredenables reader network 104 to interrupt an ongoing traversal andimmediately proceed to a new traversal. A first termination techniqueinvolves reader network 104 sending a data “NULL” symbol 502, such asthe data “NULL” symbol 502 described above with reference to FIG. 5. Asshown in the state diagram of FIG. 12A, the receipt of a data “NULL”symbol 502 causes tag 102 to enter dormant state 1202. Hence, receipt ofa data “NULL” symbol 502 causes operation in the flowchart of FIG. 13 topass to step 1302.

Once in dormant state 1202, reader network 104 may initiate a new binarytraversal by causing steps 1304–1309 of FIG. 13 to be performed. Inother words, reader network 104 may initiate a binary traversal bytransmitting a master reset signal, performing calibration procedureswith tag(s) 102, and transmitting a command for tag(s) 102 to enter treetraversal state 1208.

In a second termination technique, tags may implicitly place themselvesinto dormant state 1202 using optional step 1322. In this embodiment,tags 102 will automatically enter their dormant state 1202 aftertransmitting the last bit of data to reader network 104, as indicated instep 1322. Hence, using this termination technique, reader network 104reads bits from tags 102 until they stop responding. Hence, readernetwork 104 can read one or more tags 102 having variable data lengthswithout having prior knowledge of their variable data lengths. Once tags102 no longer respond, reader network 104 knows all tags 102 have beenread regardless of their respective data lengths.

Reader network 104 may utilize additional termination techniques torelatively quickly eliminate one or more subsets of a population of tags120. For example, subsets of a population of tags 120 may be defined bya classification number. The classification number may be located withinthe first bits of the ID number of each tag 102. A particularclassification number may be identified by a traversal that onlytraverses the bit pattern corresponding to the particular classificationnumber. Once such a distinct bit pattern is identified, reader network104 may terminate its current traversal by issuing a data “NULL.” Thesubset of tags 102 matching the classification number can then beeliminated from the current tag population search by reader network 104.

The subset of tags 102 may be eliminated as follows: By issuing the data“NULL,” all tags 102 that are assigned the particular classificationnumber would be in state 1208 (binary traversal) after addressing thesefew bits of the distinct bit pattern. Tags 102 that did not match thisparticular classification number would have at some point during thetraversal followed steps 1318, 1319, 1320, and 1321 into ‘mute’ state1212. Thus, when reader network 104 issues the data “NULL,” tags 102 instate 1208 will implicitly place themselves into dormant state 1202 byleaving step 1316 via the ‘Yes’ branch, passing to step 1302. Tags 102in dormant state 1202 will remain dormant until the next master reset isissued. When the remaining tags 102 receive the data ‘null,’ they willfollow the ‘yes’ branch from step 1321 to step 1307. Hence, they will bere-initialized to start another traversal with the first bit of theirID, which in the current embodiment is the first bit of their particularclassification number. Hence, using the ability of tags 102 to actimplicitly, reader network 104 may relatively quickly remove specificpopulations of tags from responding to traversals until the next masterreset signal.

In a similar termination technique, reader network 104 may choose toaddress a subset of the population of tags 120, ignoring other tags thatmay respond. In the preceding example, tags 102 are each assignedclassification numbers in their identification numbers, and the bits ofthese are first transmitted to the reader network 104. Reader network104 may direct a binary traversal along a path such that tags 102 havinga particular classification will follow steps 1302–1318, 1322, and 1324,and back to step 1310. Tags 102 that match the classification numberwill be in state 1208, or in binary traversal. Tags 102 that do notmatch the classification number will follow steps 1302–1318, at somepoint not matching a bit sent by reader network 104. Hence, operation ofthese tags 102 will pass to steps 1319, 1320, and 1321, where tags 102are in mute state 1212. However, reader network 104 may elect not todisable these tags 102 as in the preceding termination example, butinstead may continue following a binary traversal. However, in thisexample, only the subset of tags 102 that match the classificationnumber actively respond, and will remain in state 1208. Tags 102 that donot match the classification number will be in mute state 1212. Thesetags 102 will not respond to the reader network 104 until the next data“NULL,” as shown in the ‘yes’ branch of step 1321 that passes to step1307. In this manner, reader network 104 may specifically address asubset of the population of tags matching the particular classificationnumber. Reader network 104 ignores the responses of those tags 102having a different classification number. The ignored tags 102 will notenter a dormant state 1202 as in the preceding example, but instead willremain in mute state 1212.

For example, this ability to identify subsets of tags 102 may be appliedto identify classes of objects to which tags 102 are attached. In anexample application involving retail inventory, items belonging to aparticular class of goods (e.g., jeans, CD players, overnight shipments,etc.) are each affixed with a tag 102 having a bit pattern in itsidentification number. The bit pattern uniquely corresponds to the classof goods. Reader network 104 may identify whether items of one or moreof these particular classes exist by using a binary traversal algorithmto determine whether the particular bit patterns of the classificationsexist in the identification numbers of these tags 102. Note that inembodiments, variations in the bit pattern may correspond to differentlevels in a package hierarchy. The bit pattern variations may be used todistinguish, for example, an item tag, a box tag, a carton tag, a totetag, and a pallet tag, etc., from each other. Thus, a reader network 104could, for example, read only the pallet tag even if the pallet containsnumerous cartons of numerous boxed items that have corresponding tags.

In addition to identifying subsets of tags 102, reader network 104 mayuse termination techniques to prevent the reading of additionalinformation appended to a tag identification number. For instance, tags102 may employ tag identification numbers that include an identifyingportion and a sensor data portion, as shown in FIG. 11. The identifyingportion identifies a tag 102. However, the sensor data portion providesinformation generated by a sensor within tag 102. If reader network 104only needs identifying information, then reader network 104 mayterminate interrogations before receiving the sensor data portion. Forexample, reader network 104 may issue a data “NULL” after receivingidentification information but before receiving sensor information.

Note that FIGS. 12A and 13 do not indicate the exact bit lengths thatreader network 104 collects or that tags 102 transmit. In a preferredembodiment, the binary traversal protocol of the present inventionallows for a variable length protocol. Reader network 104 begins abinary traversal by issuing signals that step one or more tags 102through steps 1302–1318, 1322, 1324, and back to step 1310. Aspreviously noted, at any time during this process, reader network 104may issue a data ‘null,’ which transitions tag 102 into dormant state1202 via step 1316 using the ‘yes’ branch. The data “NULL” can betransmitted at any bit in the full sequence of bits of an identificationnumber. In a preferred embodiment (which does not include step 1322),tags 102 will continue to send out bits after all bits of itsidentification number have been sent and received by reader network 104.For example, without additional information to send, tag 102 willtransmit logical “0” backscatter symbols. When tag 102 includes sensorinformation, such as shown in FIG. 11, sensor information bits that areavailable at that time are transmitted after the identification number,after which “0” backscatter symbols are transmitted. Accordingly, readernetwork 104 controls the number of bits collected, which ultimatelydetermines the bit length of the population. In an embodiment, readernetwork 104 may know that tags 102 of a particular classification have aparticular bit length of identification. Reader network 104 candetermine “on the fly” when to stop reading identification bits andissue a data “NULL,” so that it can collect the next tag 102, which mayhave a different identification number length. Hence, upward mobilityfor tags 102 having longer ID numbers is present, and is an advantage ofthe present invention.

Reader network 104 may employ the binary traversal protocol tointerrogate a population of tags according to various techniques. Afirst example interrogation technique involves reading every tag 102 ina tag population that can be detected. This technique is referred toherein as a general read interrogation. During a general readinterrogation, reader network 104 traverses through the tag populationby exchanging symbols with the tag population. During this process, whenreader network 104 receives two backscatter symbols simultaneously (suchas a logical “0” and a logical “1”) in response to a transmitted signal,it selects one of these symbols to transmit next. In doing so, thereader network 104 evoke responses from any tags 102 that match thetransmitted symbol, and implicitly places the remaining, non-responsiveundesired tags 102 into mute state 1212 shown in FIG. 12A. This maycontinue until no more responses are evoked from tags 102, or apredetermined number of bits have been traversed, or until readernetwork 104 has otherwise determined it has finished traversing tags102. Embodiments for general read interrogations are described ingreater detail below with reference to FIGS. 15A–B. Note that aspects ofthe algorithms shown in FIGS. 15A and 15B may be combined.

Another interrogation technique verifies that a particular tag 102exists within its communications range. This technique is referred toherein as a specific read interrogation. During a specific readinterrogation, reader network traverses though the tag population usinga particular bit pattern. For example, the particular bit patternmatches the identification number of a particular tag 102. A preferredembodiment for a specific read interrogation technique is described ingreater detail below with reference to FIG. 14A. FIG. 14B illustrates analternative embodiment for the specific read interrogation technique.

Note that aspects of the algorithms shown in FIGS. 14A and 14B may becombined. For example, steps 1408, 1409, 1498 of FIG. 14B may beimplemented into FIG. 14A. These steps would validate proper signalsfrom the tag population on each bit, such that the algorithm of FIG. 14Awould immediately exit when an expected symbol from the target tag isnot received. This combined algorithm facilitates a faster exit from thealgorithm when the target tag is not present.

Note that the general read and specific read interrogation techniquesare similar. As described herein, reader network 104 determines whichbit values (i.e., “0” or “1”) it chooses to acknowledge. Thus, for thegeneral and specific read interrogation techniques, reader network 104controls which of tags 102 remain in tree traversal state 1208, andwhich tags 102 implicitly move to dormant state 1202.

Reader network 104 may also employ an interrogation technique that is ahybrid of the specific read and general read interrogation techniques.The hybrid technique is referred to herein as a group readinterrogation. Group read interrogations enable reader network 104 toidentify a predetermined subset of tags 102 within a tag population, ifthe predetermined subset exists.

When conducting a group read interrogation, reader network 104 initiallyperforms a specific read operation. However, the specific read operationis conducted only for a partial predetermined sequence of tagidentification bits. If one or more tags 102 respond to the partialpredefined sequence, then reader network 104 continues the group readinterrogation by performing a general read interrogation on theremaining tag identification bits. In this fashion, reader network 104addresses only a particular subset of tags 102. This is accomplished byselectively ignoring the responses of the rest of the population of tags120. This is different than specifically addressing and disablingparticular subsets of the population of tags 120. The protocolimplemented by tags 102 allows specific and non-specific tag addressing,without modification of tag 102, or modification of the manner in whichthe identification number is established or programmed into tag 102.

Further details on general read and specific read interrogations areprovided in the sub-sections below for purposes of illustration, and notlimitation. The invention is not limited to the particular examples ofcomponents and methods described herein. Alternatives (includingequivalents, extensions, variations, deviations, etc., of thosedescribed herein) will be apparent to persons skilled in the relevantart(s) based on the teachings contained herein. Such alternatives fallwithin the scope and spirit of the present invention.

3.1.1 Specific Read Interrogation Protocol Embodiments

In a preferred embodiment, the approach that reader network 104 uses toisolate and determine that a particular tag 102 exists is detailed inFIG. 14A. The process involves two stages. In the first stage, the fullidentification number of a particular chosen tag 102 is sent out.Because this identification number is unique, all other tags 102 thatare within the communications range of reader network 104 are implicitlydirected to enter into the mute state 1212. These tags 102 no longercommunicate with reader network 104. In the second stage, a pattern isacquired from the chosen tag 102. Because the chosen tag 102 is the onlyone of the population of tags 120 that can be in tree traversal state1208, if reader network 104 detects a valid pattern from a tag, thechosen tag 102 exists. Otherwise, if no valid pattern is detected,reader network 104 presumes that the chosen tag 102 does not existwithin its communications range. Note that this embodiment has anadvantage of greater noise immunity because a greater number of bits areacquired and verified from the tag.

The example steps shown in the flowchart of FIG. 14A will now bedescribed in detail. The flowchart of FIG. 14A begins with step 1401. Instep 1401, reader network 104 starts its processing by receiving a tagidentification number from a host system. Reader network 104 is directedto validate the identification number.

In step 1402, reader network 104 activates all tags 102 within itscommunications range with a master rest signal.

In step 1402 a, reader network 104 calibrates the tags 102.

In step 1403, reader network 104 issues a command to place tags 102 intotree traversal state 1208.

In steps 1404–1407, reader network 104 transmits each bit of theidentification number received by the host system until all bits aretransmitted. In a preferred embodiment, reader network 104 is notlistening or paying attention to signals received from the tagpopulation 102 during these steps. Reader network 104 merely insuresthat all but the intended tag 102 (if present) are implicitlytransitioned into mute state 1212. This completes the first stage of thealgorithm.

In step 1410, reader network 104 clears an accumulator buffer to receivea pattern of bits/symbols from tag 102.

In step 1408, a backscatter symbol is received from the population oftags 120 as a result of the last iteration of steps 1405 and 1406.

In steps 1411–1413, and passing back to step 1408, the accumulatorreceives a predetermined number of bits from the receiver section ofreader network 104 that were received from tag 102. These bits aredesirably a transmission of a pre-defined bit pattern from tag 102 thatmatches the desired identification number requested by the host system.The match is verified in the following steps:

In step 1414, after receiving the bits, reader network 104 compares thebits received from tag 102 and stored in the accumulator with thedesired pre-defined bit pattern. If the bit patterns do not match, thepresumption is false, and sought-after tag 102 is presumed not to existwithin range. This pattern of bits that were received may actually havebeen noise or were noise affected. In either case, the desired tag wasnot verified in the current environment. Control then passes to step1498. If the bit patterns do match, control passes to step 1499.

In step 1498, the host system is notified by reader network 104 that thedesired tag was not verified. The process then ends.

In step 1499, the host system is notified by reader network 104 that thedesired tag was verified. The assumption is that the bits received byreader network 104 in steps 1408 and 1411–1413 were from a valid tagwith valid signals overriding any noise in the environment. Thisassumption has a chance of error, of course, and is determinedstatistically upon the randomness of noise. As such, the longer the bitpattern, the less chance that noise created this pattern. For example, asingle-bit pattern may be considered to have a 1 in 2 chance of beinggenerated by noise, which is not generally acceptable in industry. An8-bit pattern may be considered to have a 1 in 256 (i.e., 2⁸) chance ofbeing randomly generated by noise, and so on. Choosing patterns that donot have just 0's or just 1's, which can be the pattern generated bysystematic noise, can eliminate systematic noise as a factor in thepattern.

In an alternative embodiment for the just described algorithm forverifying the existence of a particular tag, reader network 104 canreceive information from the population of tags 120 at the same time itis transmitting the identification bits of the particular desired tag102. FIG. 14B shows a flowchart providing example steps for thisapproach, according to an embodiment of the present invention. Theflowchart of FIG. 14B is described as follows:

In step 1401, similarly to the just described algorithm, reader network104 receives the identification number from a host system.

In steps 1402 and 1402 a, similarly to the just described algorithm, thepopulation of tags 120 are calibrated by sending reset andsynchronization pulses in steps 1402, 1402 a.

In step 1403, reader network 104 places tags 102 into tree traversalstate 1208.

In step 1404, reader network 104 selects the first bit to be sent fromthe working register. Reader network 104 enters the process loop ofsteps 1405, 1408, 1049, 1406, and 1407. Each bit of the desiredidentification number is processed during each trip around the loop.

In step 1405, reader network 104 transmits the next bit in theidentification number.

In step 1408, reader network 104 receives a backscatter symbol bit fromthe tag population.

In step 1409, unlike the process shown in FIG. 14A, it is determinedwhether the desired bit signal is received from the tag populationduring step 1408. If the bit signal is not received, operation proceedsto step 1498. Otherwise, reader network 104 prepares to process the nextbit, and operation proceeds to step 1406.

In step 1498, a determination is made that the desired tag does notexist, and the determination is reported to the host system.

In step 1406, it is determined whether there are any more tagidentification number bits in the working register. If there are no morebits, operation proceeds to step 1499. If there are more bits, operationproceeds to step 1407.

In step 1499, it is presumed that the desired tag exists, and thispresumption is reported to the host system.

In step 1407, reader network 104 designates the value of the next bit inits working register as the next bit of the desired identificationnumber. Operation proceeds to step 1405.

The process shown in the flowchart of FIG. 14B does not require a bitpattern to be returned to reader network 104 by the desired tag 102.However, this process may be more affected by RF noise in theenvironment.

3.1.2 General Read Interrogation Protocol Embodiments

A difference between a specific read interrogation and a general readinterrogation is that for specific read interrogations, reader network104 responds to pre-selected “0”s' and “1”s'. In contrast, for generalread interrogations, reader network 104 responds to received bits fromtags 102 according to a particular preference for the received bitsignals. The preference may be determined by a variety of differentalgorithms. In a preferred embodiment, reader network 104 has apreference for a stronger received signal. An example of this embodimentis illustrated in a flowchart in FIG. 15A. In an alternative, readernetwork 104 has a preference based on a particular signal or bit value.An example of this embodiment is illustrated in a flowchart in FIG. 15B.

In a preferred embodiment for a general read interrogation, readernetwork 104 has a preference for the strongest (i.e., relatively higheramplitude) received signal, as illustrated in FIG. 15A. For example,during a general read interrogation, reader network 104 checks todetermine whether received signals are strong enough (i.e., high enoughamplitude) to consider as valid responses from a population of tags 120.Reader network 104 then checks to determine whether the strongestreceived signal represents a “0” or a “1” logical value. An advantage ofthis approach is that reader network 104 traverses the strongestreceived signals first, and causes tags 102 that transmit thesestrongest signals to transition into dormant state 1202 after readingtheir identification numbers. Hence, while reading the population oftags 120, the first of tags 102 that are read will be less affected bynoise (i.e., due to their stronger transmitted signal strength), thusincreasing the efficiency of reader network 104. As reader network 104reads and eliminates the tags 102 transmitting stronger signals, andworks its way towards tags 102 transmitting weaker signal, the effect ofnoise on the received signals may begin to increase. In a preferredembodiment, the noise can be detected using a cyclic redundancy check(CRC) code, which is a protocol common to data communicationsindustries. Using the CRC code to detect noise as opposed to a valid tagresponse affords a reader network 104 the opportunity to force tags 102that may currently be in tree traversal state 1208 to transition intomute state 1212. These tags may then be cycled back to command state1206 by the use of a data “NULL” symbol 502, without disabling any oftags 102 that have not yet completely transmitted their entireidentification number. Note that the protocol of the present inventionallows a reader network 104 to receive a single symbol, e.g., thestrongest, as in FIG. 15A, or all symbols simultaneously, as in FIG.15B.

The embodiment shown in FIG. 15B is similar to that shown in FIG. 15A.However, there is a difference in which binary signal reader network 104will prefer to receive from the population of tags 120. This preferenceleads to the response of reader network 104. For example, during ageneral read interrogation, reader network 104 may have a preference for“0.” Thus, if reader network 104 receives a “0” from one or more tags102, reader network 104 responds by transmitting a “0.” This responsecauses tags 102 that have transmitted a “1” to reader network 104 totransition to mute state 1212. For example, this transition may occurwhen operation passes from step 1318 to step 1319, as shown in FIG. 13.Note that the protocol may alternatively be adapted to prefer a “1.”

As described above, reader network 104 may use various terminationtechniques to interrupt an ongoing traversal. Upon such a termination,reader network 104 may immediately proceed to a subsequent traversal.For instance, reader network 104 may terminate a general readinterrogation after one or more subsets of identification numbers aredetermined to exist in a population of tags 120.

For example, if each of a plurality of tags 102 in a tag population hasan identification number having 130 bits, reader network 104 may performa general read interrogation on only 10 of the 130 bits. In performingsuch an interrogation, reader network 104 may not determine exactlywhich unique tags 102 are in range. However, in performing such aninterrogation, reader network 104 is capable of learning that one ormore tags 102 exist in range. For instance, reader network 104 may gainknowledge (i.e., within a few bit reads) of the existence of aparticular subset of tags 102. This may allow for useful applications,such as the identification of bits that identify special inventoryitems, including one or more express packages located within a largenumber (e.g., thousands or more) of standard delivery packages.

FIG. 15A is a flowchart illustrating an operational sequence of apreferred embodiment of a general read interrogation from theperspective of reader network 104. Thus, FIG. 15A illustrates analgorithm that reader network 104 may use to implement a general readinterrogation of all tags 102 of a tag population within itscommunication range. This algorithm demonstrates how reader network 104can retrieve identification numbers from one or more tags 102 withoutprior knowledge of particular identification numbers within the tagpopulation. Note that some steps shown in the flowchart of FIG. 15A donot necessarily have to occur in the order shown.

The flowchart of FIG. 15A begins with step 1501. In step 1501, readernetwork 104 transmits a master reset signal. The master reset signalcauses tags 102 within range to transition to calibration state 1204,shown in FIG. 12A.

In step 1502, reader network 104 calibrates tags 102. For example,reader network 104 and tags 102 undergo oscillator and data calibrationoperations, as described elsewhere herein. After completing calibration,tags 102 transition to command state 1206.

In step 1503, reader network 104 designates the first bit of a binarytraversal, a command bit, to send to tags 102. The bit directs tags 102to transition into tree traversal state 1208.

In step 1504, reader network 104 clears its working register so that tagidentification number bits may be stored therein as they are receivedfrom tags 102 during the present binary traversal. For example, one ormore bits will be received from a particular tag 102 that is currentlyunknown.

In step 1507, reader network 104 sends the designated bit to thepopulation of tags 120.

In step 1508, reader network 104 receives one or more backscatter symbolresponses from the population of tags 120. For example, the responsestransmitted by tags 102 may be transmitted in accordance with step 1310shown in FIG. 13.

After receiving the responses from tags 102, reader network 104determines which binary traversal path (e.g., a “0” or a “1”) will betaken. To make this determination, reader network 104 has a preferencefor a particular signal. For instance, in the current example, readernetwork 104 has a preference for the strongest signal. Alternatively, asillustrated in FIG. 15B and described below, reader network 104 may havea preference for a particular bit value, such as a bit “0” or a bit “1.”

In step 1511, reader network 104 checks for the existence of a validsignal in the responses received from the tag population during step1508. In the current example, reader network 104 checks for theexistence of a signal in a wide band, which may include the encoding ofbinary “0” and binary “1.” If a valid signal exists, then operationproceeds from to step 1515. If a valid signal does not exist, operationproceeds to step 1520.

The condition where a valid signal does not exist in step 1511 may occurin a number of circumstances. For example, this condition occurs whenthere are no more tags 102 to read. Also, this condition may occur if anoisy transmission environment has taken reader network 104 through aseries of bits that do not correspond to tag identification numbers ofany tags 102. The condition may additionally occur if all tags 102within the tag population have been successfully and fully read. In thiscase, each tag 102 within the tag population transitioned to dormantstate 1202.

In step 1515, reader network 104 determines whether the value of the bitreceived according to the strongest received signal is a “0.” If thereceived bit is a “0,” operation proceeds to step 1512. If the receivedbit is not a “0,” it is presumed to be a “1,” and operation proceeds tostep 1516. Note that although step 1515 as illustrated in FIG. 15A makesa decision based upon the presence or absence of a “0” bit, the decisionmay alternatively be based upon the presence or absence of a “1” bit, asboth logic symbols are transmitted by the population of tags 120 withoutinterference.

In step 1512, the reader bit is set to a “0,” and operation proceeds tostep 1518.

In step 1516, the reader bit is set to “1,” and operation proceeds tostep 1518.

In step 1518, reader network 104 accumulates the reader bit into itsworking register. During successive performances of step 1518, readernetwork 104 builds a current tag identification number bit stream in itsworking register.

In step 1520, reader network 104 determines whether there is information(i.e., bits) stored in its working register. If the working registerdoes not contain any information, it is concluded that there are no moretags 102 within the tag population to read, and operation proceeds tostep 1522. If reader network 104 determines that the working registercontains information accumulated performance of the aforementionedsteps, operation proceeds to step 1524.

In step 1522, reader network 104 informs the host system that thegeneral read interrogation operation is complete. After performance ofstep 1522, operation may stop. However, reader network 104 may performsubsequent general read interrogations by returning to step 1501.

In step 1524, reader network 104 sends the accumulated information tothe host system. During performance of step 1524, the host system mayreceive less than a full working register of information. This may occurin a noisy transmission environment. However, in the absence of suchnoise, the host should receive a fully requested identification number.However, either the host system or reader network 104 may check thenumber of bits sent to the host system to identify whether noise orother source a loss of one or more bits. In an embodiment, a partiallyfull working register may not be transmitted the host system. In such anembodiment, step 1524 would be bypassed, and operation would proceeddirectly from step 1520 to step 1526.

The passage of operation from step 1520 to step 1524 may signify thetransition of a tag 102 to mute state 1212. Such a transition may occurin an alternative algorithm after tag 102 has transmitted all of itsbits. An example of such a transition is shown in the flowchart of FIG.13, as the branch from step 1322 to step 1302.

In step 1526, reader network 104 transmits a “NULL” symbol 502. As shownin FIG. 12A, transmission of a “NULL” symbol 502 causes each tag 102 inmute state 1212 (i.e., not yet successfully read) to transition tocommand state 1206. Once in command state 1206, tags 102 are eligiblefor subsequent traversals. Performance of step 1526 causes a differentresult for any fully read tag(s) 102 that remained in tree traversalstate 1208. As shown in FIG. 12A, receipt of the “NULL” symbol 502causes any such tags 102 to enter dormant state 1202 (i.e., as alsodescribed with respect to the tag algorithm shown in FIG. 13, withoutexecuting optional step 1322). Thus, “NULL” symbol 502 issued by readernetwork 104 during step 1526 is an implicit command, a command that isinterpreted differently depending upon the current state of each tag102.

Accordingly, in a preferred embodiment, during general readinterrogations, the responding tag population becomes smaller as tags102 are successively read and identified. The fully read and identifiedtags 102 transition into dormant state 1202. This process continuesuntil all responsive tags 102 in the tag population are identified andtransition into dormant state 1202.

After each time that step 1526 is performed, operation proceeds to step1503, where reader network 104 begins a next binary traversal by causingany tags 102 in command state 1206 to transition to tree traversal state1208.

FIG. 15B illustrates flowchart providing a procedure for reader network104 to determine which received signal type is preferred from thepopulation of tags 120, according to an alternative embodiment of thepresent invention. The algorithm shown in FIG. 15B is similar to that ofFIG. 15A, with the exception of steps 1508, 1510, and 1514.

In the example of FIG. 15B, the algorithm takes a preference for abinary symbol “0” from the population of tags 120. This is shown in FIG.15B, where reader network 104 leaves step 1508 with one or more tag 102backscatter symbol responses, as opposed to the algorithm shown in FIG.15A, where reader network 104 leaves step 1508 with at most a singlesignal stored. Note that the algorithm alternatively may have apreference for a binary symbol “1.”

In step 1510 of FIG. 15B, reader network 104 determines whether a binarysymbol “0” was received in the responses of step 1508. If this symboldoes exist, operation proceeds to step 1512. If a symbol “0” does notexist, operation proceeds to step 1514.

In step 1512, reader network 104 sets the reader bit to the “0” bit, andoperation proceeds to step 1518.

In step 1514, the reader network 104 determines whether a symbol “1” wasreceived in the responses of step 1508. If this symbol does exist, thereader network 104, operation proceeds to step 1516. However, if symbol“1” does not exist, operation proceeds to step 1520.

In step 1516, reader network 104 sets the reader bit to the “1” bit, andoperation proceeds to step 1518.

For a description of the remaining steps, refer to the description aboverelated to these steps in FIG. 15A. These example algorithms demonstratemany approaches to the control of reader network 104 over the populationof tags. In fact, many different algorithms are applicable to readernetwork 104, that allow communication with a population of tags 120, andare compatible with algorithm described above in reference to FIG. 13.According to embodiments of the present invention, a variety of readernetworks may be implemented that balance different degrees of costs andabilities to read tags 102 in a noisy environment, all while beingcompatible with the same tags 102 In other words, tags 102 do notrequire modification to be compatible with different embodiments ofreader network 104, according to the present invention.

3.2 Traversal of an Exemplary Tag Population

FIG. 16 is an illustration an exemplary population of tags 120, thatincludes a first, a second, and a third tag 102 a, 102 b, and 102 c.FIG. 16 shows the traversal of tags 102 a–c in a binary tree format. Forexemplary purposes, each tag has a third bit long identification number.The binary tree shown in FIG. 16 has three levels, where each levelcorresponds to a bit in the three bit identification number. The firstlevel of the binary tree, is the start level, which corresponds to steps1401–1404 shown in FIG. 14A, and to steps 1501–1504 shown in FIG. 15A.The second level of the binary tree represents the first bit of theidentification numbers of tags 102 a, 102 b, and 102 c (reading from theleft). The second level corresponds to steps 1405 and 1507 as shown inFIGS. 14 and 15, respectively.

In FIG. 16, the “0” branch of the binary tree descends towards the left,and the “1” branch descends towards the right. As described above withreference to FIG. 15A, reader network 104 may prefer a particular signalwhen performing a general read interrogation. For example, as shown inthe flowchart of FIG. 15B, reader network 104 prefers “0” (i.e.,descending towards the left in FIG. 16) on any combination of signalsreceived. Each successive downward level in the binary tree diagramrepresents another bit read from tags 102. Each branch in the binarytree diagram represents a decision (i.e., a command) and a bittransmitted by reader network 104.

A first example shows how reader network 104 reads tag 102 b through aspecific read interrogation technique. For exemplary purposes, the bitsare not inverted into a least significant bit (LSB) first format.Instead, for ease of illustration, the examples based on FIG. 16 flowfrom most significant bit (MSB) to LSB, as read from left to right. Theoperational sequence of FIG. 14B is followed. First, reader network 104receives “100” from a host system, where the host system desires toconfirm the existence of tag 102 b. Reader network 104 performs steps1401–1405 shown in FIG. 14A, and transmits bit “0” to tags 102, which isan instruction to enter tree traversal state 1208. In FIG. 16, theseoperations occur at point 1605.

At this point, tags 102 a, 102 b, and 102 c are active, because thesetags have reached step 1309 of FIG. 13. Tags 102 a, 102 b, and 102 cthen send their first bit, pursuant to step 1310. Accordingly, tag 102 atransmits a “0” backscatter symbol 602 and tags 102 b and 102 c transmit“1” backscatter symbols 802.

Due to receiving the symbols from tags 102 a–c, reader network 104 hasreceived a plurality of signals (e.g., a “0” backscatter symbol and two“1” backscatter symbols). Operation proceeds through step 1409 (on theyes branch) as a bit “1” was received. Reader network 104 requiresadditional bits to be read, and thus operation proceeds through step1406 to step 1407, which designates its target bit of “1” (i.e., thefirst bit of “100” received from the host) as the reader bit. Because,reader network 104 finds the target bit “1” in the received combinationsignal, this bit is transmitted to tags 102 a–c, pursuant to step 1405.In FIG. 16, this is illustrated as a move down the logical “1” path frompoint 1605 to point 1606. Point 1606 represents storage of a bit “1.”

Tags 102 a–c receive the “1” symbol 402 transmitted by reader network104. However, different responses occur among tags 102 a–c. The bit “0”transmitted by tag 102 a does not match the bit sent from reader network104. Accordingly tag 102 a transitions to mute state 1212, as shown asthe path from 1318 to step 1319 in FIG. 13. Tag 102 a now effectivelyawaits a data “NULL” signal, which would indicate a new binary traversalby reader network 104.

Since tags 102 b and 102 c each transmitted a “1” backscatter symbolthat matched the bit transmitted by reader network 104, tags 102 b and102 c load their next bit to be transmitted. Tag 102 b loads a “0” bit,and tag 102 c loads a “1” bit). Tags 102 b and 102 c transmit these bitsas backscatter signals 702 and 902, respectively, to reader network 104pursuant to steps 1316, 1318, 1324 and 1310. Reader network 104 loadsthe second bit of the tag identification number, “0,” (step 1407), andreceives a plurality of signals in step 1408 (i.e., the transmitted “0”and “1” backscatter symbols). In steps 1409 and 1406, reader network 104determines that a “0” bit was received, and that there are additionalbits to collect. Hence, operation of reader network 104 proceeds to step1405, and transmits the “0” symbol 302. In FIG. 16, the transmission ofthe “0” symbol 302 by reader network 104 is illustrated as a move topoint 1607. Hence, point 1607 represents receiving the bits of “10.”

Tag 102 c receives the “0” symbol 302 transmitted by reader network 104(step 1312). However, tag 102 c had last transmitted a “1” backscattersymbol (step 1310). Hence, because these bits do not match (step 1318),tag 102 c transitions to mute state 1212 (step 1319), and thereforeawaits the next binary traversal.

Tag 102 b receives the “0” symbol 302, and because it had lasttransmitted a “0” backscatter symbol, the bits do match (step 1318).Operation proceeds to step 1310, where tag 102 b transmits its next bitof “0” as backscatter symbol 602.

Reader network 104 loads the next target bit of “0” (step 1407), andreceives the bit “0” transmitted by tag 102 b (step 1408). These bitsmatch, and operation therefore proceeds from step 1409 to step 1406.Reader network 104 determines that all 3 bits of the identificationnumber are received, in step 1406. Reader network 104 may now report thematch to the host system (step 1499). This result is illustrated in FIG.16 as point 1608, where reader network 104 has stored “100.” Thus,reader network 104 has successfully determined the existence of tag 102b.

In another example described as follows, reader network 104 reads alltags 102 in range, without prior knowledge of their existence. Thus, theexample describes a general read interrogation. For this example, readernetwork 104 operates according to the flowchart shown in FIG. 15A, andtags 102 operate according to the flowchart shown in FIG. 13. Each passof algorithm shown in FIG. 15A selects the strongest tag signal from theremaining members of the tag population. For this example, we willassume that tags 102 a, 102 b, and 102 c are in the order of strongestto weakest transmitted signals.

The general read interrogation example begins with reader network 104performing steps 1501–1504. In step 1507, reader network 104 selects alogical “0” bit as the reader bit and transmits this value to tags 102a–c. In FIG. 16, the steps are represented by point 1601.

Tags 102 a, 102 b and 102 c receive this first transmitted reader bitfrom reader network 104, pursuant to step 1309. Pursuant to step 1310,each of tags 102 a–c designates and sends their first identificationnumber bit to reader network 104. Hence, reader network 104 receives aplurality of signals of a “0” backscatter symbol 602 (sent by tag 102a), and “1” backscatter symbols 802 (sent by tags 102 b and 102 c). Inthe exemplary flowchart of FIG. 15A, reader network 104 receives/selectsthe strongest signal in step 1508, which in the current example is fromtag 102 a. Because the received signal is a “0” backscatter symbol,reader network 104 stores the “0” bit value in its working register.Furthermore, reader network 104 transmits the “0” symbol 302, pursuantto steps 1511, 1515, 1512, 1518, 1519, and 1507.

Tags 102 a, 102 b and 102 c receive the transmitted “0” symbol 302,pursuant to step 1312. However, in performing step 1318, tags 102 b and102 c determine that this received bit does not match the bit they havemost recently sent. Therefore, operation of tags 102 b and 102 cproceeds to step 1319, where they each transition to mute state 1212,and wait for the next binary traversal operation. In FIG. 16, theseoperations are represented by point 1602.

Unlike tags 102 b and 102 c, tag 102 a determines (by performing step1318) that the received “0” symbol 302 matches the prior bit transmittedby tag 102 a. Therefore, pursuant to steps 1324 and 1310, tag 102 adesignates a next bit of its identification number (i.e., a “1”) andsends this bit as a “1” backscatter symbol 802 to reader network 104.

Therefore, reader network 104 receives a single “1” backscatter symbolfrom the tag population (i.e., because tag 102 b and 102 c are in thenon-transmitting mute state 1212). With reference to FIG. 15A, operationof reader network 104 proceeds from step 1511 upon the receipt of avalid signal, to step 1515. Because a “0” backscatter symbol was notreceived, operation of reader network 104 proceeds to step 1516, wherethe reader bit is set to the received “1” bit. Reader network 104accumulates the reader bit in its working register (which now contains“01”), and transmits this bit “1” to tags 102 a–c. In FIG. 16, theseoperations are represented by point 1603.

Tag 102 a receives the transmitted “1” symbol 402, and by performingstep 1318, determines that it matches the bit value most recentlytransmitted to reader network 104. Therefore, pursuant to steps 1324 and1310, tag 102 a designates the next bit of its identification number(i.e., a “1”) and sends this designated bit as a “1” backscatter symbol902 to reader network 104.

Reader network 104 receives the transmitted “1” backscatter symbol andperforms step 1511, step 1515, and step 1516, where it determines that a“1” bit has been received. Reader network 104 accumulates this bit inits working register (which resultantly stores “011”). This isrepresented by point 1604 shown in FIG. 16. Hence, reader network 104collected all bits of the identification number of tag 102 a, andoperation proceeds to step 1524. In step 1524, the identification numberof tag 102 a is sent to the host system.

As described above with reference to FIG. 13, tags 102 may performalternative steps when bits are matched in step 1318. In one suchalternative embodiment, tags 102 perform an optional step 1322. If tag102 a executes step 1322, it will determine that all of itsidentification number bits have been transmitted to reader network 104.Accordingly, operation of tag 102 a proceeds to step 1302, where tag 102a transitions to dormant state 1202. As a result, none of tags 102 a–cis operating in tree traversal state 1208. Therefore, reader network 104receives no response from tags 102 a–c. As a result, operation of readernetwork 104 proceeds to step 1520 where a determination is made that theworking register of reader network 104 is not empty, because it hasaccumulated the identification number “011.” In step 1524, readernetwork 104 sends this identification number to the host system.

If tag 102 a does not perform optional step 1322, operation of tag 102 aproceeds from step 1318 to step 1324, where a next bit of itsidentification number is designated for transmission. Because tag 102 ahas transmitted all of its identification number bits, it may designatean arbitrary bit. For instance, an arbitrary bit may be selectedaccording to register rotation techniques employed by tag 102 a duringthe performance of step 1324.

At this point, reader network 104 has completed the first binarytraversal of the general read interrogation. Reader network 104 performsstep 1526 by transmitting a “NULL” symbol 502. This “NULL” symbol 502causes all tags that are in mute state 1212 to transition to commandstate 1206.

Pursuant to reader network 104 having transmitted the “NULL” symbol 502,the next binary traversal begins, represented by point 1605 in FIG. 16.Reader network 104, pursuant to steps 1503, 1504, and 1507, transmits asignal that causes tags 102 b and 102 c to transition from command state1206 to tree traversal state 1208. Tag 102 a, however, remains indormant state 1202. Note that while reader network 104 performs step1504, it clears its working register to allow accumulation of the nexttag ID.

Tags 102 b and 102 c each send their first ID bit, which in this exampleis a “1” bit. Accordingly, reader network 104 receives a “1” backscattersymbol in step 1508, executes steps 1511 and 1515, and branches to step1516. In step 1518, the “1” bit is accumulated into its workingregister. The “1” symbol 402 is transmitted by reader network 104 instep 1507. In FIG. 16, this interrogation process portion is illustratedas point 1606.

In response to the transmission of the “1” symbol 402, tags 102 b and102 c each transmit their next identification number bit to readernetwork 104. Tag 102 b transmits a “0” backscatter symbol 702 and tag102 c transmits a “1” backscatter symbol 902. As tag 102 b's signal isstronger, reader network 104 executes steps 1508, 1511, 1515, 1512, andstep 1518, where reader network 104 sets the reader bit to “0.” In step1507, reader network 104 transmits the “0” symbol 302 to tags 102 a–c.This interrogation process portion is shown in FIG. 16 as point 1607.

After receipt of the “0” symbol 302, in step 1318, tag 102 c determinesthat the received bit does not match the bit tag 102 c previouslytransmitted. Thus, according to step 1319, tag 102 c enters mute state1212. However, tag 102 b determines that the received “0” symbol 302matches the bit tag 102 b previously transmitted. Tag 102 b executessteps 1318, 1324, and 1310, and transmits its last identification numberbit (a “0” bit) as a “0” backscatter symbol 602 to reader network 104.

Reader network 104 receives the “0” backscatter symbol 602 from tag 102b and, in accordance with step 1518, accumulates the “0” bit into itsworking register. Thus, the working register stores the binary value of“100.” Accordingly, in step 1519, reader network 104 determines that ithas accumulated a complete tag identification number. Therefore,according to steps 1524 and 1526, reader network 104 transmits thestored identification number to the host system, and transmits a “NULL”symbol 502 to tags 102 a, 102 b, and 102 c. The “NULL” symbol 502transitions tag 102 b to dormant state 1202 and tag 102 c to commandstate 1206. Tag 102 a remains in dormant state 1202.

Tag 102 c is the final tag, and therefore is traversed next, in the samemanner as the binary traversals that identified tags 102 a and 102 b.During this binary traversal, reader network 104 only receives andre-transmits bits transmitted by tag 102 c, because tag 102 c is theonly tag in tree traversal state 1208. Accordingly, with reference toFIG. 16, reader network 104 will traverse through points 1609, 1610,1611, and 1612. Upon reaching point 1612, reader network 104 (throughperformance of step 1519) determines that a complete tag identificationnumber has been accumulated. In step 1524, reader network 104 transmitsthe identification number to the host system. Reader network 104transmits a “NULL” symbol 502, which causes tag 102 c to transition todormant state 1202.

After transmitting the “NULL” symbol 502, reader network 104 performssteps 1503, 1504, 1507, and 1508. However, because tags 102 a–102 c areeach in dormant state 1202, no responses are received in step 1508.Therefore, operation of reader network 104 passes through step 1511 tostep 1520. In step 1520, reader network 104 determines that theaccumulator is empty. Operation proceeds to step 1522, where readernetwork 104 informs the host system that the general read interrogationis complete.

Through the above-described example general read interrogation, readernetwork 104 determined the existence of three previously unknown tags,tags 102 a–c, in an efficient manner. More particularly, reader network104 performed only three binary traversals to collect the identificationnumbers of these tags.

As described above with reference to FIG. 15A, reader network 104gathers identification number bits from a particular tag 102 until itdetermines (e.g., in step 1519) that it has accumulated a completeidentification number. To support tag populations employing differentsize identification numbers, reader network 104 may adjust the number ofreceived bits it requires to recognize a complete identification number.

In the example of binary traversal described above with reference toFIG. 16, reader network 104 collected identification number bits indecreasing order of significance. That is, the most significant bit(MSB) was retrieved first, and the least significant bit (LSB) wasretrieved last. However, it is within the scope and spirit of thepresent invention to retrieve bits in any order of significance during abinary traversal. For example, bits may be retrieved in an increasingorder of significance during a binary traversal. Retrieval of bits fromone or more tags 102 in this order is useful for interrogating tagpopulations where one or more of the higher significant identificationnumber bits are not used.

For example, consider a tag population where a seven-bit longidentification number is used. In this population, there are three tags102 having the following respective identification numbers: “0000011,”“0000100,” and “0000110.” The four MSBs in each of these identificationnumbers is “0000.” By determining this bit pattern characteristic of theidentification number, reader network 104 may bypass retrieval of thefour MSBs during a binary traversal, and use just the three LSBs touniquely identify the tags 102. This bypass feature further streamlinesinterrogation operations. With reference to FIG. 15A, reader network 104may implement this bypass feature in step 1519. For this exemplary tagpopulation, reader network 104 may indicate in step 1519 that a completeidentification number has been accumulated after the collection of justthree out of the original seven bits.

To facilitate the bypass feature, many methods can be implemented. Theseinclude, but are not limited to: a single pre-scan by a reader network104; multiple pre-scans by the reader network 104, and; an algorithmicapproach based upon previous general read interrogations in particularcircumstances calculated by the host system, where the host systemprovides instructions on how to perform the bypass operation to readernetwork 104. In a preferred embodiment of the bypass feature, readernetwork 104 causes a population of tags 120 to transition intosuperposition state 1210, and exchanges signals (i.e., performs a scanof) with the population of tags 120. During the exchange of signals,reader network 104 determines a range of identification numbers thatexist in the population of tags 120. Accordingly, FIGS. 17A and 17B showflowcharts that illustrate this determination from the perspective of aparticular tag 102 and a particular reader network 104, respectively.

3.3 Superposition Mode Communication Embodiments

FIG. 17A shows a flowchart illustrating operation of a tag 102 insuperposition state 1210. As shown in FIG. 17A, operation begins withstep 1702. In step 1702, tag 102 is in dormant state 1202.

In step 1704, tag 102 receives a master reset signal from reader network104. Upon receipt of this signal, tag 102 transitions from dormant state1202 to calibration state 1204.

In step 1706, tag 102 is synchronized with reader network 104.Accordingly, oscillator calibration and data calibration proceduresoccur, that are further described below. After tag 102 becomessynchronized with reader network 104, it enters command state 1206.

In step 1708, tag 102 receives a command from reader network 104 thatcauses tag 102 to transition to superposition state 1210. As describedabove, the command may be a one or more bits, such as a single bitlogical “1” symbol 402.

In step 1710, tag 102 designates an initial bit of its identificationnumber for transmission as a backscatter symbol to reader network 104.This designated bit may be any bit of the identification number. Forexample, this designated bit may be either the MSB or the LSB of theidentification number. This bit chosen assumes that each tag 102 in thetag population is encoded in the same manner, where tags 102 allimplement MSB to LSB or all implement LSB to MSB. Operation of tag 102proceeds to a loop that begins with step 1712.

In step 1712, tag 102 receives a symbol from reader network 104.

In step 1714, tag 102 determines whether the symbol received in step1713 is a “NULL” bit. If the received symbol is a “NULL” bit, operationof tag 102 proceeds to step 1720. If the received symbol is not a “NULL”bit, operation of tag 102 proceeds to step 1718.

In step 1718, tag 102 determines whether the received bit matches thedesignated identification number bit. If the bits do not match,operation proceeds to step 1722. If the bits do match, operationproceeds to step 1719.

In step 1719, tag 102 sends the designated bit as a backscatter symbolto reader network 104. Operation proceeds to step 1722.

In step 1720, tag 102 transitions to command state 1206

In step 1722, tag 102 designates a next bit of its identification numberfor transmission to reader network 104. The next bit may be designatedin any number of ways. For instance, if the initial bit designated instep 1710 is the LSB, then tag 102 may designate the identificationnumber bit as the bit having the next highest significant bit position.Alternatively, if the initial bit designated bit in step 1710 is theMSB, then tag 102 may designate the identification number bit as the bithaving the next lowest significant bit position. With reference to theexemplary tag architecture shown in FIG. 10, these features may beimplemented in state machine 1024 through various register rotationtechniques. After step 1722, operation of tag 102 proceeds to step 1712for the next iteration of the loop. Tag 102 will exit the loop afterreceiving a “NULL” symbol 502 from reader network 104, and willtransition to a command state 1206 via step 1720.

As described above, tag 102 eventually returns to command state 1206 instep 1720. From this step, tag 102 may enter tree traversal state 1208.Thus, operation of tag 102 may proceed from step 1720 to step 1308 inFIG. 13.

Thus, the flowchart of FIG. 17A shows that while in superposition state1210, a population of tags 120 provide feedback to reader network 104regarding the existence of identification number bits in the population.For example, by sending a stream of “0” symbols 302, reader network 104can determine whether identification numbers containing a “0” bit ateach particular transmitted bit position exist in the population of tags120. In addition, reader network 104 can determine the position in theidentification number of each of the these bits. This feature enablesreader network 104 to determine an identification number rangeassociated with a tag population. In essence, reader network 104performs a logical bit wise “or” of the signals of the entire populationof tags 120.

Accordingly, FIG. 17B shows a flowchart illustrating operation of areader network 104 while information is being collected from apopulation of tags 120. Operation begins with step 1750. In step 1750,reader network 104 causes each tag in the population of tags 120 totransition into superposition mode 1210. As shown in FIG. 17B, step 1750includes steps 1752 and 1753.

In step 1752, reader network 104 transmits a master reset signal, whichcauses all tags 102 in the population of tags 120 to transition tocalibration state 1204. Reader network 104 engages in calibrationprocedure(s) with tags 102. These procedures may include oscillator anddata calibration operations, as described herein.

In step 1753, reader network 104 transmits a command, such as a singlebit, that causes each tag 102 in the population of tags 120 to entersuperposition state 1210.

In step 1760, reader network 104 determines the position(s) of “0” bitsin the identification numbers of the population of tags 120. FIG. 17Bshows that step 1760 includes steps 1762–1768.

In step 1762, reader network 104 designates an initial bit position.

In step 1764, reader network 104 transmits a “0” symbol 302.

In step 1765, reader network 104 determines whether any “0” backscattersymbols were received in response from the population of tags 120. Ifone or more “0” backscatter symbols were received, operation proceeds tostep 1766

In step 1766, reader network 104 marks the designated bit position ascontaining one or more “0” bits before returning control to step 1767.

In step 1767, reader network 104 determines whether it has designatedall identification number bit positions. If all identification numberbit positions have been designated, operation proceeds to step 1770. Ifall identification number bit positions have not been designated,operation proceeds to step 1768

In step 1768, reader network 104 designates a next identification numberbit. Operation proceeds to step 1764 to complete the processing loop.

In step 1770, reader network 104 ensures that the population of tags 120is again initialized in superposition mode 1210, so that tags 102 willeach designate their initial identification number bit. Thus, step 1770may include the step where reader network 104 transmits a “NULL” signalto cause each tag 102 to transition to command state 1206, and transmitsa command that transitions each tag 102 into superposition mode 1210.However, step 1770 is optional. For example, in embodiments where tags102 perform circular register rotation techniques to designate andtransmit identification number bits, each tag 102 may be designating itsinitial identification number bit upon completion step 1760.

In step 1780, reader network 104 determines the positions of “1” bits inthe tag population's identification numbers, in a manner very similar tothat described above for step 1760. As shown in FIG. 17B, step 1780includes steps 1782–1788.

In step 1782, reader network 104 designates an initial bit position.

In step 1784, reader network 104 transmits a “1” symbol 402.

In step 1785, reader network 104 determines whether it received any “1”backscatter symbols in response to the “1” symbol 402 transmitted instep 1784. If a “1” backscatter symbol was received, operation proceedsto step 1786.

In step 1786, reader network 104 marks the designated bit position ascontaining one or more “1” bits, and operation proceeds to step 1787.

In step 1787, reader network 104 determines whether it has designatedall identification number bit positions. If all identification numberbit positions have been designated, operation proceeds to step 1790. Ifall identification number bit positions have not been designated,operation proceeds to step 1788

In step 1788, reader network 104 designates a next identification numberbit. After step 1788, operation proceeds to step 1784 to complete theprocessing loop.

In step 1790, reader network 104 determines the number of identificationnumber bits required for interrogation. This step includes readernetwork 104 identifying the last read bit position (in the sequence ofbit positions that reader network 104 receives tag identification numberbits), where every identification number bit in the tag populationcollectively has multiple values, both “0” and “1.”

For example, the determination step 1790 may include reader network 104first collecting the LSB of an identification number, and proceeding tocollect subsequent adjacent identification number bits in increasingorder of significance. In an example, reader network 104 interrogates apopulation of tags 120 that have four 7-bit identification numbers:0000100, 0000010, 0000111, and 0000101. In performing steps 1760 and1780, reader network 104 determines that, for these four identificationnumbers, the four MSB positions include a single bit value of “0.”

Therefore, reader network 104 identifies the third LSB position as thelast bit position (in the sequence of bit positions that reader network104 receives tag identification number bits) where every identificationnumber bit in the population of tags collectively multiple values (i.e.,“0” and “1” bit values). Accordingly, reader network 104 determines thatonly three bits need to be collected to uniquely identify tags 102 inthis population. Reader network 104 stores the single signal values ofthe remaining 4 bits, hereby designated as the superposition mask. Thus,with reference to the flowchart shown in FIG. 15A, reader network 104determines in step 1519 that a complete tag identification number isknown after only three bits have been collected. Thus, reader network104 can provide to the host system a complete tag ID in about 3/7 of thetime required when the superposition function is not used.

A similar result occurs when reader network 104 interrogates a tagpopulation having four 7-bit identification numbers: 1010100, 1010010,1010111, and 1010101. In performing steps 1754 and 1758, reader network104 determines that, for these identification numbers, the four mostsignificant bit positions of the four identification numbers contain thesame bit pattern of “1010.” Accordingly, reader network 104 determinesthat only three bits must be collected to uniquely identify tags 102 inthis population. Thus, as in the prior example, reader network 104determines in step 1519 that a complete tag identification number isknown after only the first three bits are collected.

As described above, reader network 104 may collect bits in any order.Accordingly, reader network 104 may also employ the techniques of FIG.17B for any such order of bit collection. For example, reader network104 may first collect an identification number's MSB and proceed tocollect adjacent bits in decreasing order of significance. For each ofthese cases, reader network 104 may collect fewer than all of theinterrogation bits when one or more of the LSBs are the same for theentire population of tags 120. According to the present invention,reader network 104 may collect bits in any sequence of “0” and “1” bits.

4. Timing Subsystem Embodiments of the Present Invention

4.1 Timing Subsystem Overview

Structure and operation of timing subsystem 1023 shown in FIG. 10 isfurther described in this section. Timing subsystem 1023 provides systemclocking and data timing functions for tag 102. As described below,timing subsystem 1023 provides a system clock for integrated circuit1002. Timing subsystem 1023 also provides frequencies used by RFinterface portion 1021 to generate backscatter modulated symbols. Timingsubsystem 1023 also provides for oscillator calibration and for datacalibration. These functions are further described below.

In the embodiment shown in FIG. 10, timing subsystem 1023 includes SAR1022, state machine 1024, oscillator 1026, counter 1028, first divider1036, and second divider 1038. Tag 102 of the present invention usesoscillator 1026 to serve as a system time reference for internal digitalfunctions. Two additional frequencies are obtained from oscillator 1026to be used for encoding data to be transmitted from tag 102, using firstand second dividers 1036 and 1038. SAR 1022 is used during oscillatorcalibration. Counter 1028 is used for oscillator calibration, datacalibration, and data timing. Example embodiments for oscillator 1026are described in the next sub-section, followed by a description of acalibration procedure for oscillator 1026, and a description of a datacalibration procedure.

4.2 Oscillator Configurations

The present invention requires an accurate oscillator signal to be usedto control the operation of logic circuitry. The oscillator signal mayalso be used to produce two or more data frequencies for transmittedsignals. For example, according to the present invention, a firstfrequency is used as a data frequency for transmitted “1” data bits. Asecond frequency is used as a data frequency for transmitted “0” databits.

A benefit in having a relatively tight tolerance range for the sourceoscillator frequency is that it provides for relatively tight tolerancesfor the transmitted data frequencies from one tag 102 to another tag 102in the population of tags 120. The frequency spectrum ranges fortransmitted “1”s and “0”s from the population of tags 120 cannotoverlap, or even be too close, or they may be confused for each other bya reader network 104. By increasing the accuracy of the sourceoscillator frequency, the respective frequency bands for transmitted“1”'s and “0”'s are narrower and therefore can be closer togetherwithout overlap. Furthermore, with narrower frequency bands, eachfrequency band can be closer to the carrier frequency without overlap.Hence, lower frequencies may be used, which can lead to less overallpower consumption. Hence, the ability to calibrate the oscillatorfrequency such that it has a relatively tight tolerance range isdesirable.

Crystal oscillators are very accurate, and may be used in someimplementations for oscillator 1026. However, crystal oscillators arerelatively large, expensive, and may not be practical for use in a smallspace. Preferably, oscillator 1026 is implemented as an oscillatorcircuit in a semiconductor technology such as CMOS. In this manner,oscillator 1026 may be incorporated “on chip” with other portions of thecircuitry of the present invention, taking up relatively little area.Furthermore, CMOS is widely available and relatively inexpensive tomanufacture. However, CMOS process variations can cause such anoscillator to have a frequency variation of +−50% from CMOS chip to CMOSchip.

In a preferred embodiment, the oscillator of the present invention is acircuit implemented in CMOS. FIG. 18 shows an example adjustableoscillator 1026, according to an embodiment of the present invention.Oscillator 1026 receives a control word 1070 of a length of one or morebits, and outputs a master clock signal 1062. The frequency of masterclock signal 1062 is determined by a base internal frequency ofoscillator 1026, and by control word 1070. Adjustable oscillator 1026outputs an oscillator frequency on master clock signal 1062 that isequal to the base internal frequency adjusted according to control word1070. Hence, adjustable oscillator 1026 outputs an oscillator frequencyon master clock signal 1062 that may be adjusted upward and/or downwardaccording to control word 1070.

FIG. 19 shows an oscillator configuration that provides for multipleoscillator frequencies, according to an embodiment of the presentinvention. The oscillator configuration shown in FIG. 19 includesadjustable oscillator 1026, first divider 1036, and second divider 1038.First divider 1036 is a divide-by-three divider. Second divider 1038 isa divide-by-two divider. In an embodiment, adjustable oscillator 1026outputs a frequency of 7.5 MHz on master clock signal 1062. Firstdivider 1036 receives master clock signal 1062, and divides thefrequency of master clock signal 1062 by 3. When master clock signal1062 has a frequency of 7.5 MHz, first divider 1036 outputs a firstclock signal 1066 having a frequency of 2.5 MHz. Second divider 1038receives master clock signal 1062, and divides the frequency of masterclock signal 1062 by 2. When master clock signal 1062 has a frequency of7.5 MHz, second divider 1038 outputs a second clock signal 1064 having afrequency of 3.75 MHz. Hence, three frequencies are provided by theoscillator configuration of FIG. 19: 2.5 MHz, 3.75 MHz, and 7.5 MHz. Theselection of these frequencies, according to a preferred embodiment,prevents harmonics from the 2.5 MHz band from intruding into the 3.75MHz band, which could cause errors during symbol detection by readernetwork 104. Note that these frequency values are provided for purposesof illustration. The present invention is applicable to any suitableoutput frequency for oscillator 1026, and to alternative division valuesfor first and second dividers 1036 and 1038.

Adjustable oscillator 1026 may be implemented in any number ofoscillator circuit configurations, including resistor-capacitor (RC)oscillator and ring oscillator configurations. RC oscillator, ringoscillator, and additional oscillator configurations that are adaptableto the present invention are well known to persons skilled in therelevant art(s). For illustrative purposes, an example RC oscillator isdescribed at a high level as follows. In a RC oscillator circuitimplementation, the oscillator frequency is determined by the values ofone or more resistors and capacitors. The values of one or more of theresistors and/or capacitors may be altered to change the oscillatorfrequency. FIG. 20 illustrates an example block diagram of an RCoscillator implementation for adjustable oscillator 1026, according toan embodiment of the present invention. Adjustable oscillator 1026includes a reference logic 2002, a feedback logic 2004, a frequencyadjustment bank 2006, and a comparator 2008.

Comparator 2008 generates master clock signal 1062. Comparator 2008compares the two signals at its inputs: a reference signal 2010 and afrequency adjustment signal 2012. If frequency adjustment signal 2012 isgreater than reference signal 2010, comparator 2008 will output alogical low value for master clock signal 1062. If frequency adjustmentsignal 2012 is less than reference signal 2010, comparator 2008 willoutput a logical high value for master clock signal 1062.

Reference logic 2002 generates a relatively stable reference voltagethat is output on reference signal 2010. Reference logic 2002 mayinclude whatever passive or active elements are required to generate thereference, including transistors, resistors, capacitors, inductors, andamplifiers. The voltage value for the reference voltage is selected asrequired by the particular application.

Frequency adjustment bank 2006 includes a bank of one or more frequencyadjustment elements that are switchable by corresponding bits of n-bitcontrol word 1070. Frequency adjustment bank 2006 typically includes abase frequency adjustment element, used to determine a base frequencyfor adjustable oscillator 1026. The base frequency adjustment elementmay include one or more of capacitors and resistors used for at least aportion of the RC time constant for the base frequency of the RCoscillator implementation. Each additional element of the bank offrequency adjustment elements includes one or more resistors and/orcapacitors that may be switched in parallel or series with the basefrequency adjustment element to alter the base frequency. A switchcontrolled by a bit of n-bit control word 1070 may be used to switch ina particular frequency adjustment element. Frequency adjustment bank2014 outputs a frequency adjustment signal 2012.

Feedback logic 2004 receives master clock signal 1062 and frequencyadjustment signal 2012 from frequency adjustment bank 2006. Feedbacklogic 2004 includes one or more logical, active, and passive componentsto condition master clock signal 1062 as necessary. Feedback logic 2004may include one or more capacitors that form a portion of the R-C timeconstant for the base frequency of the RC oscillator implementation.Feedback logic 2004 couples frequency adjustment signal 2012 to masterclock signal 1062, so that frequency adjustment signal 2012 will rampupwards and downwards depending on whether master clock signal 1062 iscurrently a high or a low logical level. Frequency adjustment signal2012 will ramp upwards and downwards at a rate controlled by the currentR-C time constant determined by frequency adjustment bank 2006 andfeedback logic 2004.

When master clock signal 1062 is low, frequency adjustment signal 2012will ramp downward until it ramps below the level of reference signal2010. At this point, comparator 2008 will change its output to a highlevel. Frequency adjustment signal 2012 will then ramp upwards until isramps above the level of reference signal 2010. When this happens,comparator 2008 will change its output to a low level, repeating theprocess. In this manner, master clock signal 1062 is an oscillatingsignal, and the frequency of the oscillation is controlled.

The oscillator embodiments provided above in this section are presentedherein for purposes of illustration, and not limitation. The inventionis not limited to the particular examples of components and methodsdescribed herein. Alternatives (including equivalents, extensions,variations, deviations, etc., of those described herein) will beapparent to persons skilled in the relevant art(s) based on theteachings contained herein. Such alternatives fall within the scope andspirit of the present invention.

4.3 Oscillator Calibration

Variations in manufacturing and fabrication processes can causevariations in semiconductor characteristics that affect operation.Variations in semiconductor operation may occur due to variations intemperature, humidity, and other environmental factors, and due tomanufacturing process variations, etc. For example, there may bevariation between different semiconductor wafer lots, between differentwafers within a particular lot, and in different areas of a singlewafer. In CMOS, resistor and capacitor values may each have tolerancesof ±25%, due to the above described variations. In an RC oscillatorconfiguration, the combination of tolerance values can lead to anoverall oscillator frequency tolerance range of ±50%. This is arelatively large tolerance range. Hence, it is desirable for adjustableoscillator 1026 to be able to be calibrated across an oscillatorfrequency tolerance range of ±50%.

According to a conventional calibration method, the oscillator frequencymay be tested and adjusted once during the manufacturing process.However, such an adjustment accounts for process variations, notenvironmental variations. Therefore, because characteristics of theoscillator circuit may change over time due to environmental variations,the oscillator frequency may eventually drift outside an acceptabletolerance range. Hence, it would be beneficial to allow for calibrationof the oscillator frequency at one or more times subsequent tomanufacturing.

The present invention allows for calibration of the oscillator frequencydynamically, during circuit operation, as often as is needed by theparticular application. FIG. 21A shows a portion of timing subsystem1023 of FIG. 10 used for oscillator calibration, according to anembodiment of the present invention. As shown in FIG. 21A, timingsubsystem 1023 includes adjustable oscillator 1026, a successiveapproximation register (SAR) 1022, and a counter 1028. Timing subsystem1023 allows for dynamic calibration of the oscillator frequency.

Timing subsystem 1023 is used to calibrate adjustable oscillator 1026according to an input signal 2100. Input signal 2100 may be a signalthat was received “off-chip” from an integrated circuit hosting timingsubsystem 1023, in a wired or wireless fashion, or may also have beenreceived “on chip.” For example, input signal 2100 may be a data signalobtained from a signal received by tag 102. Input signal 2100 may be oneor received signals 1050 a and 1050 b shown in FIG. 10, or a processedform of received signals 1050 a and 1050 b output by state machine 1024.In embodiments, timing subsystem 1023 calibrates adjustable oscillator1026 such that for each cycle of input signal 2100, adjustableoscillator 1026 converges as close as possible to a predeterminedfrequency, measured by the number of cycles or pulses, that oscillator1026 generates during a cycle of input signal 2100. For example,adjustable oscillator 1026 may ideally output a series of 255 pulses forevery pulse received on input signal 2100. If more or less than 255pulses are output by adjustable oscillator 1026 during a cycle of inputsignal 2100, the frequency of master clock signal 1062 is adjusted. Inother words, timing subsystem 1023 calibrates oscillator 1026 to afrequency dictated by one or more signals transmitted by reader network104, regardless of what the value of that frequency is.

Counter 1028 receives input signal 2100 and master clock signal 1062.Counter 1028 is a counter or timer that counts the number of cycles ofmaster clock signal 1062 that occur during a cycle of input signal 2100.Counter 1028 outputs a count word 1074 equal to the number of cycles ofmaster clock signal 1062 that occurred during a cycle of input signal2100.

Successive approximation register (SAR) 1022 receives input signal 2100and count word 1074. SAR 1022 monitors one or more bits of count word1074. SAR 1022 alters control word 1070 if the monitored bit(s)indicates that too many or too few cycles of master clock signal 1062occur during a cycle of input signal 2100. Each bit of control word 1070may be adjusted according to a different reading of count word 1074. Forexample, SAR 1022 may successively adjust the bits of control word 1070,from highest order bit to lowest order bit, or vice versa, to adjustcontrol word 1070 to an increasingly finer degree. State machine 1024may aid in the operation of SAR 1022. As shown in FIG. 10, state machine1024 may be coupled between counter 1028 and SAR 1022. When coupledbetween counter 1028 and SAR 1022, state machine 1024 receives countword 1074 and outputs processed count word 1072, which is received bySAR 1022.

Adjustable oscillator 1026 receives the altered control word 1070 fromSAR 1022, and adjusts the frequency output on master clock signal 1062accordingly. In this manner, timing subsystem 1023 calibrates adjustableoscillator 1026. Two or more iterations that adjust count word 1074 andcorrespondingly adjust control word 1070 may be used to increasinglyfine tune the frequency output by adjustable oscillator 1026. Furtherdetails regarding oscillator calibration are provided in the followingsubsections.

FIG. 21B illustrates a more detailed block diagram of timing subsystem1023, according to an embodiment of the present invention. Thisembodiment is described in further detail as follows. In the descriptionthat follows, the base frequency for adjustable oscillator 1026 is 7.5MHz.

As shown in FIG. 21B, adjustable oscillator 1026 receives control word1070. Control word 1070 is shown as an 8 bit wide signal. FIG. 23B showsan example value for control word 1070. Each possible value for controlword 1070 directs adjustable oscillator 1026 to output a correspondingfrequency. For example, the minimum and maximum values for control word1070 vary the output frequency of adjustable oscillator 1026 by +50% and−50%, respectively, from its base frequency. When control word 1070 isequal to 00000000, oscillator 1026 outputs its base frequency plus 50%,which is 11.25 MHz. When control word 1070 is equal to 11111111,oscillator 1026 outputs its base frequency, minus 50%, which is 3.75MHz. Values for control word 1070 that are in between these causeoscillator 1026 to output corresponding frequencies in between 3.75 MHzand 11.25 MHz. For example, when control word 1070 is equal to 10000000(i.e., a middle binary value), adjustable oscillator 1026 outputs itsbase oscillator frequency on master clock signal 1062 (i.e., 7.5 MHz).

First divider 1036 is optional. When present, first divider 1036receives and divides master clock signal 1062, and outputs first clocksignal 1066. In the embodiment shown in FIG. 21B, first divider 1036 isa divide-by-3 divider. Hence, when master clock signal 1062 is afrequency of 7.5 MHz, first clock signal 1066 outputs a frequency of 2.5MHz.

Counter 1028 receives first clock signal 1066 and input signal 2100.First clock signal 1066 is used as the clock signal for the internallogic of counter 1028. Input signal 2100 is received by counter 1028.When a falling edge is received on input signal 2100, counter 1028 iscleared, such that a logical zero signal is output on count word 1074.After being cleared, counter 1028 may begin counting according to firstclock signal 1066 from the zero initial state.

FIG. 22A shows an example calibration waveform cycle for input signal2100. At time 2202, input signal 2100 goes from a logical high level toa logical low level, which clears counter 1028. Hence, at time 2202,count word 1074 is forced to a logical zero state. After input signal2100 transitions to a logical low at time 2202, counter 1028 counts fromthe zero state according to first clock signal 1066. When input signal2100 transitions from a logical high level to logical low level at time2204, count word 1074 is again cleared so that counter 1028 can againbegin counting at zero.

As shown in FIG. 22A, the time period between time 2202 and time 2204 isreferred to as a calibration signal or test 2206. One or more of testssuch as test 2206 are used to calibrate adjustable oscillator 1026,according to the present invention. Counter 1028 counts from zerostarting at time 2202 until time 2204. At time 2204, SAR 1022 uses thecount value in count word 1074 to adjust the output frequency ofadjustable oscillator 1026. After test 2206 is complete, another testmay occur to further adjust the output frequency of adjustableoscillator 1026. As many tests as are required may be used to adjust theoutput frequency of adjustable oscillator 1026 until it is within anacceptable tolerance range. For example, as shown in FIG. 22B, a seriesof eight calibration signals or tests may be used: first test 2206, asecond test 2208, a third test 2210, a fourth test 2212, a fifth test2214, a sixth test 2216, and a seventh test 2218. Each test maysuccessively adjust the frequency of adjustable oscillator 1026 to afiner degree. For example, a first test 2206 may adjust the frequency ofadjustable oscillator 1026 by 50% of the adjustable amount in onedirection. The subsequent tests may adjust the frequency of adjustableoscillator 1026 by 25%, 12.5%, 6.25%, 3.125%, 1.563%, 0.781%, and0.391%.

In an embodiment, the duration of test 2206, which the is time periodbetween falling edges on input signal 2100 at times 2202 and 2204, isideally equal to 2^(j−1)−1 cycles of first clock signal 1066, where j isthe number of stages in counter 1028. In an embodiment, j is equal to 9,and hence the time period for test 2206 is:(2^(j−1)−1)×1/f _(c1)=(2⁸−1)×1/(2.5 MHz)=255×1/(2.5 MHz)=102 μSwhere f_(c1) is equal to the desired frequency of first clock signal1066. Because the frequency of master clock signal 1062 may vary due totemperature and process variations, the number of cycles of first clocksignal 1066 that occur during this time period may be greater or lessthan 255. Hence, master clock signal 1062 will need calibration.

SAR 1022 receives one or more bits of count word 1074, and uses thereceived bit(s) to modify control word 1070. FIG. 24 shows a blockdiagram for an example SAR 1022, according to an embodiment of thepresent invention. SAR 1022 includes an n-bit register bank 2402 and astate machine 2404. The n-bit register bank 2402 stores control word1070. In an embodiment, state machine 2404 initializes and sets orresets registers in n-bit register bank 2402 according to bit 8 of countword 1074 and input signal 2100. In alternative embodiments, one or moreother bits of count word 1074 can be used by SAR 1022 in addition to, orinstead of bit 8. In an alternative embodiment, state machine 2404 is aportion of state machine 1024.

Depending on the state of one or more bits of count word 1074, statemachine 2404 adjusts one or more bits of control word 1070. As shown inFIG. 24, state machine 2404 receives bit 8 of count word 1074. Bit 8 ofcount word 1074 is an overflow bit. If bit 8 is equal to a one, thismeans that counter 1028 counted too fast, and therefore counted too highduring the last cycle of input signal 2100. Hence, first clock signal1066 would need to be slowed down. If bit 8 is equal to a zero, thismeans that counter 1028 either counted at the correct rate, or countedtoo slow, during the last cycle of input signal 2100. Hence, first clocksignal 1066 would need to maintain the same rate, or increase the rate.State machine 2404 uses bit 8 and input signal 2100 to generateset/reset signals 2406 to n-bit register 2402. In an embodiment, statemachine 2404 can set a bit of one of the registers of n-bit register2402 to decrease the frequency of master clock signal 1062, or can reseta bit to increase the frequency. In alternative embodiments, multiplebits may be set or reset in n-bit register 2402 to increase or decreasethe frequency of master clock signal 1062.

4.3.1 Embodiments for Configuring an RC Oscillator Calibration Circuit

FIG. 21C illustrates a more detailed block diagram of timing subsystem1023, according to an embodiment of the present invention. As shown inFIG. 21C, oscillator 1026 is implemented using an RC oscillator similarto the RC oscillator shown in FIG. 20. Furthermore, frequency adjustmentbank 2006 is shown in more detail, according to an example embodiment ofthe present invention. As shown in FIG. 21C, frequency adjustment bank2006 includes a first capacitor 2110, and n switchable capacitors 2112a–n, where all of the capacitors are coupled in parallel. Note that thepresent invention is applicable to alternative elements in frequencyadjustment bank 2006.

FIG. 21C also shows a switch 2180 that receives a reset signal 2182,according to an embodiment of the present invention. When present,switch 2180 may be used to reset oscillator 1026 when desired, tocontrol/synchronize the phase of master clock signal 1062. For instance,the phase of master clock signal 1062 in a particular tag 102 may besynchronized with the phase of a master clock signal located in a nearbytag 102. Reset signal 2180 may be derived from a signal received by tag102 from reader network 104. When reset signal 2180 turns off switch2180, capacitors 2110 and 2112 a–n in frequency adjustment bank 2006 areshorted to ground. When reset signal 2180 subsequently turns on switch2180, operation of oscillator 1026 begins, and master clock signal 1062is initialized—i.e., capacitors 2110 and 2112 a–n begin charging from aground potential. Hence, a signal from reader network 104 may be used tosimultaneously initialize a master clock signal in one or more tags 102within communication range. In embodiments, the signal used to generatereset signal 2180 may be a calibration pulse or data symbol transmittedby reader network 104, for example.

FIG. 21D shows additional detail for an example embodiment of frequencyadjustment bank 2006. As shown in FIG. 21D, a MOSFET switch is coupledin series with each of the n switchable capacitors shown in FIG. 21C.For example, first switch 2114 a is coupled in series with firstswitchable capacitor 2112 a, second switch 2114 b is coupled in serieswith second switchable capacitor 2112 b, and an nth switch 2114 n iscoupled in series with nth switchable capacitor 2112 n. Each switch iscontrolled by a corresponding bit of control word 1070. Thecorresponding bit of control word 1070 turns a switch on or off, torespectively switch in or out the corresponding switchable capacitor inparallel with the remaining capacitors. This creates a controlledcapacitance for the RC oscillator of oscillator 1026, to in turn adjustthe frequency output by oscillator 1026 on master clock signal 1062.Hence, a bit of control word 1070 that is a logical “1” value switchesin a capacitor 2112, and oscillator 1026 oscillates at a lower rate.Conversely, a bit of control word 1070 that is a logical “0” valueswitches out a capacitor 2112, and oscillator 1026 oscillates at ahigher rate. Note that in alternative embodiments, frequency adjustmentbank 2006 may be configured such that a bit of control word 1070 that isa logical “1” value may cause oscillator 1026 to oscillate at a higherrate, and vice versa.

A process for configuring elements of this embodiment is described infurther detail as follows. In particular, a process for determiningvalues for first capacitor 2110, for switchable capacitors 2112 a–n, andfor the number n of bits in control word 1070, is provided.

In the description below: f_(o)=the oscillator frequency of master clocksignal 1062; t_(c)=the period of a single calibration waveform sent fromreader network 104, and; N=the value of count word 1074, whereN=f_(o)t_(c). For f₀=f_(c), where f_(c) is the desired center frequency,the corresponding counter value is N_(c)=f_(c)t_(c). Note that in theexample shown in FIG. 21C, and described below, the frequency of masterclock signal 1062 is directly applied to counter 1028, instead of beingdivided, as shown in FIG. 21B. The discussion below is applicable to theoscillator frequency being divided, as would be understood by personsskilled in the relevant art(s).

The value of count word 1074, N, may be expressed as:

$N = {\sum\limits_{i = 0}^{j - 1}\;{P_{i}2^{j - 1 - i}}}$Where

-   -   j=number of bits in count word 1074 of counter 1028,    -   i=the bit number in count word 1074, where 0≦i≦j−1, and    -   P_(i)=bit value, 0 or 1, where P_(o) is the MSB, P_(j−1) is the        LSB        Hence, the value of the ith bit position in count word 1074,        N_(i), is equal to:        N _(i) =P _(i)2^(j−1−i)        The center value of count word 1074, Nc, is defined as:

${N_{c} = {{\sum\limits_{i = 1}^{j - 1}\; 2^{j - 1 - i}} = {{\sum\limits_{i = 0}^{j - 2}\; 2^{j - 2 - i}} = {2^{j - 1} - 1}}}}{\mspace{25mu}\;}$for  all   Pi = 1The maximum value of count word 1074, N_(max), is defined as:

${N_{\max} = {{\sum\limits_{i = 0}^{j - 1}\; 2^{j - 1 - i}} = {2^{j} \cong {2N_{c}}}}}{\mspace{25mu}\;}$for  all  P_(i) = 1The value of count word 1074 when the MSB=1, N₀, is defined as:

${N_{0} = {{\sum\limits_{i = 0}^{j - 1}\;{P_{i}2^{j - 1 - i}}} = {2^{j - 1} = {N_{c} + 1}}}}\mspace{31mu}$with  only  P₀ = 1, all  other  P_(i) = 0Hence, N₀ represents the value of count word 1074 incremented oncebeyond the center value of count word 1074, N_(c).

Therefore, for any oscillator frequency where f₀>f_(c), P₀=1, and forany oscillator frequency where f_(o)≦f_(c), P₀=0. Hence, the value P₀may be used to determine whether f_(o) is greater than or less thanf_(c).

The values for capacitors in frequency adjustment bank 2006 may becalculated as follows. In FIG. 21C, the oscillator frequency, f_(o), isinversely proportional to the total controlled capacitance, C_(total):

$f_{o} = \frac{a}{C_{total}}$Where:

-   -   a=a design constant.

$C_{total} = {{C_{c1} + {C_{0}{\sum\limits_{m = 0}^{n - 1}\;{b_{m}2^{n - 1 - m}}}}} = {C_{c1} + {C_{0}R}}}$

-   -   R=value stored in SAR 1022    -   n=number of stages of controlled capacitance corresponding to        the number of bit stages in SAR 1022    -   m=bit number corresponding to a capacitor stage    -   b_(m)=value of m^(th) bit, determining whether a capacitor is        either enabled (b_(m)=1) or not (b_(m)=0)    -   C₀ is a base capacitance value for the n controlled capacitors,        the value of each of the n controlled capacitors being        determined by the each term of the summation show in the above        equation.    -   C_(c1) is a fixed capacitor such that        -   C_(total)=C_(c), corresponding to f_(c)=a/C_(c), where        -   C_(c)=C_(c1)+C₀R₀=C_(c1)+C₀2^(n−1)    -   Where:

$R_{0} = {{\sum\limits_{m = 0}^{n - 1}\;{b_{m}2^{n - 1 - m}}} = 2^{n - 1}}$

-   -    for b₀=1, all other b_(m)=0    -   C_(c) is the center value of C_(total), where f_(o) would equal        f_(c) if there are no process variations requiring calibration.

Setting:

$C_{c} = \frac{a}{f_{c}}$Then the value for C_(c1) is given by:

$C_{c1} = {\frac{a}{f_{c}} - {C_{0}2^{n - 1}}}$

A maximum possible capacitance value, C_(max), for frequency adjustmentbank 2006 is configured when all b_(m)=1 in the C_(total) equation shownabove:C _(max) =C _(c1) +C ₀ R _(max) =C _(c1) +C ₀(2^(n)−1)

A minimum possible capacitance value, C_(min), for frequency adjustmentbank shown in FIG. 21C is configured when all b_(m)=0 in the C_(total)equation shown above:C_(min)=C_(c1)

A maximum capacitance range ΔC that can be accommodated by theadjustable bank of capacitors is:ΔC _(total) =C _(max) −C _(min) =C ₀(2^(n)−1)The change from C_(c) to C_(max) is:ΔC ₊ =C _(max) −C ₀ =C ₀(2^(n)−2^(n−1)−1)=C ₀2^(n−1)(2−1−2^(−n+1))≅C₀2^(n−1)The change from C_(min) to C_(c) is:ΔC ⁻ =C _(c) −C _(min) =C ₀2^(n−1)Note that:

${f = \frac{a}{C_{total}}},{f_{\max} = \frac{a}{C_{\min}}},{f_{\min} = \frac{a}{C_{\max}}},{f_{c} = \frac{a}{C_{c}}}$and${C_{c} = \frac{a}{f_{c}}},{C_{\min} = \frac{a}{f_{\max}}},{C_{\max} = \frac{a}{f_{\min}}}$whereby

${\Delta\;{C\_}} = {{C_{c} - C_{\min}} = {{\frac{a}{f_{c}} - \frac{a}{f_{\max}}} = {C_{0}2^{n - 1}}}}$Hence, C₀ may be determined as follows:

$C_{0} = {a\; 2^{1 - n}\left( {\frac{1}{f_{c}} - \frac{1}{f_{\max}}} \right)}$

Accordingly, in an embodiment, a desired precision for tuning theoscillator frequency is equal to:

${\partial p} = {\frac{\partial f}{f_{c}} = \frac{1}{2^{n}}}$$\;{{2^{n} = \frac{1}{\partial p}},{{n\mspace{14mu}\log\mspace{14mu} 2} = {{\log\mspace{14mu}\left( {f_{\max} - f_{\min}} \right)} - {\log\mspace{14mu}{\square f}}}}}$$\mspace{14mu}{n = \frac{\log\left( \frac{1}{\partial p} \right)}{\log\mspace{11mu} 2}}$

Hence, the above described methodology may be used to determinecapacitance values C_(c1) and C₀ for the capacitors of frequencyadjustment bank 2006 shown in FIG. 21C, and the value n. The presentinvention is also adaptable to alternative methodologies for configuringelements of data subsystem 1023.

As described above, FIG. 21C illustrates a switch 2180. Switch 2180causes the output signal of oscillator 1026, master clock signal 1062,to be at a known phase. Switch 2180 is controlled by phase reset signal2182. In a preferred embodiment, an edge or pulse on phase reset signal2182 is triggered by every falling edge on the input signal 2100. Thephase of the output signal of oscillator 1026 is reset at each datafalling edge on input signal 2100. Hence, the phases of all tags 102within operating range of reader network 104, such as is depicted inFIG. 1, are coordinated. Without the ability to reset the phase ofoscillator 1026, the oscillators of one or more tags 102 may eventuallybecome sufficiently out of phase such that the backscatter signals thatare generated by tags 102 become out of phase with each other. When thebackscatter signals become sufficiently out of phase, the may have thedisadvantage of canceling each other, so that the backscatter signalswill not be detected by reader network 104. Note that only somephase-critical applications may be affected by this problem, and as suchswitch 2180 is optional. In a further embodiment, as shown in FIG. 21D,a falling edge detector 2184 may also be present. Falling edge detector2184 may be used to detect a rising (or falling) edge of input signal2100 to generate phase reset signal 2182.

Switch 2180 in FIG. 21C may be used to reset the exemplary RC oscillatorcircuit shown in FIGS. 20 and 21C. Alternative circuits may be used toperform this function in alternative configurations for oscillator 1023without departing from the spirit and scope of the present invention.

4.3.2 Operational Embodiments for Oscillator Calibration

Exemplary operational embodiments are presented in this section (and itssubsections). The methods are presented herein for purposes ofillustration, and not limitation. The invention is not limited to theparticular examples of components and methods described herein.Alternatives (including equivalents, extensions, variations, deviations,etc., of those described herein) will be apparent to persons skilled inthe relevant art(s) based on the teachings contained herein. Suchalternatives fall within the scope and spirit of the present invention.

In the following discussion, a series of eight calibration cycles ortests are performed on adjustable oscillator 1026. The eight bits ofcontrol word 1070 stored in n-bit register bank 2402 are checkedone-by-one and potentially altered, in order from the highest order bitto the lowest order bit. In effect, the frequency of master clock signal1062 is checked and altered by successively smaller frequency amounts,until it is within an acceptable tolerance range.

Operation of calibration circuit 2102 shown in FIG. 21B is described asfollows. Count word 1074 is shown as a 9-bit wide signal (i.e., bits 0through 8) and control word 1070 is an 8 bit wide signal. An example 9bit value for count word 1074 is shown in FIG. 23A. SAR 1022 uses bit 8of count word 1074 to determine whether an adjustment of the oscillatorfrequency is necessary. For example, if the value of bit 8 of count word1074 is equal to a first state (i.e., a logical “1”), a bit of controlword 1070 is set. If the value of bit 8 of count word 102 is equal to asecond state (i.e., a logical “0”), a bit of control word 1070 is reset.The bit of control word 1070 that is selected to be set or reset dependson the amount of adjustment of the oscillator frequency required. Inembodiments, a series of tests are performed that adjust the oscillatorfrequency according to an increasingly finer amount until it is withinthe desired tolerance range. Eight tests are performed, as shown in theexample of FIG. 22B, and are described as follows with respect to FIG.25D and shown below in Table 1:

TABLE 1 Bit 8 value SAR end value SAR start value Of count (control wordTest (control word 1070) word 1074 1070) 2206 10000000 1 10000000 220811000000 0 10000000 2210 10100000 0 10000000 2212 10010000 1 100100002214 10011000 0 10010000 2216 10010100 1 10010100 2218 10010110 110010110 2220 10010111 0 10010110

The first column of Table 1 indicates which test is being performed fora particular row. The second column of Table 1 shows the value forcontrol word 1070 at the beginning of the test (for example, as set bysteps 2522 and 2524 shown in FIG. 25D, and further described below). Thethird column of Table 1 shows example values for bit 0, P₀, (MSB) incount word 1074 received at the end of each test. The fourth column ofTable 1 shows the corresponding change in control word 1070 aftercompletion of that row's test (for example, as set by steps 2528 and2530 shown in FIG. 25D, and further described below).

Prior to the beginning of test 2206, SAR 1022 is initialized, such thatcontrol word 1070 is the 8-bit word of 10000000 (as shown in the secondcolumn of Table 1 and determined by steps 2522 and 2524 shown in FIG.25D, and further described below, for example). This value of controlword 1070 is targeted to cause adjustable oscillator 1026 to output abase oscillator frequency (i.e., 7.5 MHz) from which it can be adjusted.In the current example, the tolerance range for the base oscillatorfrequency of adjustable oscillator 1026 is +50%. By increasing ordecreasing the value of control word 1070, the frequency of master clocksignal 1062 may be correspondingly increased or decreased.

At time 2202, test 2206 is initiated by the falling edge of input signal2100. Counter 1028 begins incrementing count word 1074 from a zero stateaccording to first clock signal 1066. At time 2204, SAR 1022 receivesthe value of bit 0, P₀, (MSB) of count word 1074. If bit 0 of count word1074 is a 1, this indicates that counter 1028 is counting too fast, andhence adjustable oscillator 1026 must be operating at too high of afrequency. Bit 0 (MSB) of control word 1070 would be kept at a 1 value,to keep the frequency of adjustable oscillator 1026 the same. If bit 0of count word 1074 is a 0, this indicates that counter 1028 is countingat the proper rate, or too slowly. Bit 0 of control word 1070 would thenbe reset to a 0 value to increase the frequency of adjustable oscillator1026. As shown in Table 1, for test 2206, bit 0 of count word 1074 isequal to a 1. When bit 0 is equal to a 1, this indicates that adjustableoscillator 1026 is operating too fast. Hence, SAR 1022 leaves bit 0 ofcontrol word 1070 in a 1 state, as shown in column 4 of Table 1.

At time 2204, test 2208 is initiated by the falling edge of input signal2100. For test 2208, SAR 1022 retains the value for control word 1070created during test 2206, and additionally sets bit 1 of control word1070 to a logical high, according to operations 2534,2524. Setting bit 1of control word 1070 causes adjustable oscillator 1026 to decrease thefrequency of master clock signal 1062 (i.e., the frequency is decreasedby half of the amount of the frequency achieved of the previousadjustment). Counter 1028 clears count word 1074, and then beginsincrementing count word 1074 according to first clock signal 1066. Attime 2222, SAR 1022 reads the value of bit 0 of count word 1074. Asshown in Table 1, for test 2208, bit 0 of count word 1074 is equal to a0. When bit 0 is equal to a 0, this indicates that adjustable oscillator1026 is operating at the correct rate, or too slow. Hence, SAR 1022resets bit 1 of control word 1070 to a 0 state, as shown in column 4 ofTable 1, to cause adjustable oscillator 1026 to speed up.

At time 2222, test 2210 is initiated by the falling edge of input signal2100. For test 2210, SAR 1022 retains the value for control word 1070created by test 2208, and additionally sets bit 2 of control word 1070to a logical high. Setting bit 2 of control word 1070 causes adjustableoscillator 1026 to decrease the frequency of master clock signal 1062(although the frequency is decreased by half of the amount of theprevious adjustment). Counter 1028 clears count word 1074, and thenbegins incrementing count word 1074 according to first clock signal1066. At time 2224, SAR 1022 reads the value of bit 0 of count word1074. As shown in Table 1, for test 2210, bit 0 of count word 1074 isequal to a 0. When bit 0 is equal to a 0, this indicates that adjustableoscillator 1026 is operating at the correct rate, or too slow. Hence,SAR 1022 resets bit 2 of control word 1070 to a 0 state, as shown incolumn 4 of Table 1, to cause adjustable oscillator 1026 speed up.

At time 2224, test 2212 is initiated by the falling edge of input signal2100. For test 2212, SAR 1022 retains the value for control word 1070created by test 2210, and additionally sets bit 3 of control word 1070to a logical high. Setting bit 3 of control word 1070 causes adjustableoscillator 1026 to decrease the frequency of master clock signal 1062(although the frequency is decreased by half of the amount of theprevious adjustment). Counter 1028 clears count word 1074, and thenbegins incrementing count word 1074 according to first clock signal1066. At time 2226, SAR 1022 reads the value of bit 0 of count word1074. As shown in Table 1, for test 2212, bit 0 of count word 1074 isequal to a 1. When bit 0 is equal to a 1, this indicates that adjustableoscillator 1026 is operating too fast. Hence, SAR 1022 leaves bit 3 ofcontrol word 1070 in a 1 state, as shown in column 4 of Table 1, to keepadjustable oscillator 1026 at the tested frequency.

At time 2226, test 2214 is initiated by the falling edge of input signal2100. For test 2214, SAR 1022 retains the value for control word 1070created by test 2212, and additionally sets bit 4 of control word 1070to a logical high. Setting bit 4 of control word 1070 causes adjustableoscillator 1026 to decrease the frequency of master clock signal 1062(although the frequency is decreased by half of the amount of theprevious adjustment). Counter 1028 clears count word 1074, and thenbegins incrementing count word 1074 according to first clock signal1066. At time 2228, SAR 1022 reads the value of bit 0 of count word1074. As shown in Table 1, for test 2214, bit 0 of count word 1074 isequal to a 0. When bit 0 is equal to a 0, this indicates that adjustableoscillator 1026 is operating at the correct rate, or too slow. Hence,SAR 1022 resets bit 4 of control word 1070 to a 0 state, as shown incolumn 4 of Table 1, to cause adjustable oscillator 1026 to speed up.

At time 2228, test 2216 is initiated by the falling edge of input signal2100. For test 2216, SAR 1022 retains the value for control word 1070created by test 2214, and additionally sets bit 5 of control word 1070to a logical high. Setting bit 5 of control word 1070 causes adjustableoscillator 1026 to decrease the frequency of master clock signal 1062(although the frequency is decreased by half of the amount of theprevious adjustment). Counter 1028 clears count word 1074, and thenbegins incrementing count word 1074 according to first clock signal1066. At time 2230, SAR 1022 reads the value of bit 0 of count word1074. As shown in Table 1, for test 2216, bit 0 of count word 1074 isequal to a 1. When bit 0 is equal to a 1, this indicates that adjustableoscillator 1026 is operating too fast. Hence, SAR 1022 leaves bit 5 ofcontrol word 1070 in a 1 state, as shown in column 4 of Table 1, to keepadjustable oscillator 1026 at the tested frequency.

At time 2230, test 2218 is initiated by the falling edge of input signal2100. For test 2218, SAR 1022 retains the value for control word 1070created by test 2216, and additionally sets bit 6 of control word 1070to a logical high. Setting bit 6 of control word 1070 causes adjustableoscillator 1026 to decrease the frequency of master clock signal 1062(although the frequency is decreased by half of the amount of theprevious adjustment). Counter 1028 clears count word 1074, and thenbegins incrementing count word 1074 according to first clock signal1066. At time 2232, SAR 1022 reads the value of bit 0 of count word1074. As shown in Table 1, for test 2218, bit 0 of count word 1074 isequal to a 1. When bit 0 is equal to a 1, this indicates that adjustableoscillator 1026 is operating too fast. Hence, SAR 1022 leaves bit 6 ofcontrol word 1070 in a 1 state, as shown in column 4 of Table 1, to keepadjustable oscillator 1026 at the tested frequency.

At time 2232, test 2220 is initiated by the falling edge of input signal2100. For test 2220, SAR 1022 retains the value for control word 1070created by test 2218, and additionally sets bit 7 (LSB) of control word1070 to a logical high. Setting bit 7 of control word 1070 causesadjustable oscillator 1026 to decrease the frequency of master clocksignal 1062 (although the frequency is decreased by half of the amountof the previous adjustment). Counter 1028 clears count word 1074, andthen begins incrementing count word 1074 according to first clock signal1066. At time 2234, SAR 1022 reads the value of bit 0 of count word1074. As shown in Table 1, for test 2220, bit 0 of count word 1074 isequal to a 0. When bit 0 is equal to a 0, this indicates that adjustableoscillator 1026 is operating at the correct rate, or too slow. Hence,SAR 1022 resets bit 7 of control word 1070 to a 0 state, as shown incolumn 4 of Table 1, to cause adjustable oscillator 1026 to increase toits final adjusted value.

After test 2220, the calibration sequence is complete, and the value forcontrol word 1070 shown in column 4 of Table 1 for test 2220 is thevalue selected to continue to control the frequency for adjustableoscillator 1026, until the next calibration sequence. Note thatadjustable oscillator 1026 may be calibrated at any time, as required bythe particular application. For example, oscillator 1026 may becalibrated each time that tag 102 is reset.

Hence, calibration circuit 2102 iteratively adjusts the frequency outputby adjustable oscillator 1026 on master clock signal 1062 until it iswithin an acceptable tolerance range. Master clock signal 1062 may beadjusted by this calibration process over a range of 2^(n)−1 values,wherein n is the width of control word 1070 and the number of tests oriterations. When control word 1070 is 8 bits wide, master clock signal1062 may be adjusted over a range of 2⁸−1 values, or 255 values. Forexample, master clock signal 1062 may be adjusted from a base frequencyupwards by 127 values, and downwards by 128 values. When the basefrequency is equal to 7.5 MHz, and the tolerance range is ±50%, the basefrequency of 7.5 MHz may be adjusted ±3.75 MHz, or over a span of 7.5MHz. Hence, the base frequency may be adjusted upwards and downwards inincrements of 7.5 MHz/255=29.4 KHz. This potentially leads to atolerance range for master clock signal 1062 after calibration of29.4 KHz/7.5 MHz×100%=0.39%.Note that in some environments, worst case noise estimates couldeffectively negate the last bit or bits of calibration.

Note that not all available bits of control word 1070 must necessarilybe tested during the above described calibration routine. Inembodiments, a subset of the available bits of control word 1070 may bepermanently pre-set during manufacturing or fabrication of the circuit.For example, circuits within a wafer may be tested during manufacturing.This can determine variations that will tend to occur across the wafer,that can be calibrated out. Bits may be pre-set by a variety of knownprocesses, such as by hardwiring, by pre-programming, by laser make-linkor break-link, by blowing traces, and by other known means. This may beaccomplished in SAR 1022, oscillator 1026, or on the signal traces ofcontrol word 1070. By pre-setting one or more of the available bits ofcontrol word 1070, time may be saved during calibration, because thecalibration routine will not need to test all available bits.

The calibration circuit embodiments provided above in this section arepresented herein for purposes of illustration, and not limitation. Forexample, the invention is applicable to alternative bit widths forcontrol word 1070 and count word 1074, to different frequencies thanthose discussed, and to different polarities of bits for count word 1074and control word 1070, as would be understood by persons skilled in therelevant art(s) from the teachings herein. The invention is alsoapplicable to alternative implementations for SAR 1022 than shown inFIG. 24. The invention is not limited to the particular examples ofcomponents and methods described herein. Alternatives (includingequivalents, extensions, variations, deviations, etc., of thosedescribed herein) will be apparent to persons skilled in the relevantart(s) based on the teachings contained herein. Such alternatives fallwithin the scope and spirit of the present invention.

Furthermore, note that alternative calibration waveforms may be used,having alternative polarities, duty cycles, and additional cycles. Forexample, FIG. 29 shows a calibration or test waveform, test 2900, thatmay be used alternatively to the calibration waveforms described above,such as test 2206, to calibrate adjustable oscillator 1026. Test 2900includes a calibration waveform cycle 2902, similar to that of test2206. Furthermore, test 2900 includes a separation pulse 2908 thatfollows calibration waveform cycle 2902. Separation pulse 2908 may beused to provide separation between calibration waveform cycle 2902 andthe subsequent calibration waveform, so that SAR 1022 and adjustableoscillator 1026 have time to adjust the oscillator frequency of masterclock signal 1062 before the next calibration pulse. Separation pulse2908 may be of any applicable length and duty cycle, including 3 μS highand 3 μS low.

FIG. 25A shows a flowchart 2500 providing steps for calibrating anoscillator frequency with an input signal, according to embodiments ofthe present invention. FIGS. 25B–C provide steps according to furtherembodiments. The steps of FIGS. 25A–C do not necessarily have to occurin the order shown, as will be apparent to persons skilled in therelevant art(s) based on the teachings herein. Other structuralembodiments will be apparent to persons skilled in the relevant art(s)based on the following discussion. These steps are described in detailbelow.

In the embodiments according to flowchart 2500, the oscillator frequencyis calibrated according to an input signal. For example, the oscillatorfrequency is the frequency of master clock signal 1062, and the inputsignal may be input signal 2100. A clock signal is equal to theoscillator frequency divided by an integer amount. For example, theclock signal is clock signal 1066, which is generated from master clocksignal 1062 by first divider 1036.

Flowchart 2500 begins with step 2502. In step 2502, a count word isincremented after each cycle of the clock signal that occurs during acalibration cycle of the input signal. For example, the count word iscount word 1074, as shown in FIG. 23A, and output by counter 1028.Counter 1028 increments count word 1074 each cycle of clock signal 1066.Counter 1028 increments count word 1074 during a particular test, suchas test 2206 shown in FIG. 22A, which is a calibration cycle waveform ofinput signal 2100.

In step 2504, the oscillator frequency is adjusted based upon the countword after completion of step 2502. For example, as shown in FIGS. 21Aand 21B, SAR 1022 receives count word 1074, and outputs a control word1070, which is used to adjust the output frequency of adjustableoscillator 1026.

For example, in an embodiment, step 2504 may include the step where theoscillator frequency is based on a control word. In other words, theoutput of adjustable oscillator 1026, master clock signal 1062, is basedupon control word 1070. For example, adjustable oscillator 1026 outputsa base oscillator frequency, such as 7.5 MHz, for a middle value forcontrol word 1070, such as 10000000. As control word 1070 is variedupward and downward, the frequency of adjustable oscillator 1026 willlikewise vary. For example, the frequency of master clock signal 1062may be varied by ±50% from the base frequency of 7.5 MHz.

In an embodiment, step 2504 may include the step where a bit of thecontrol word is adjusted based upon the count word. For example, Asdescribed above, a bit of control word 1070 is adjusted during eachcalibration cycle, or test. The bit of control word 1070 is adjustedaccording to the value of count word 1074 in the example discussionprovided above.

FIG. 25B illustrates additional steps for flowchart 2500, according tofurther embodiments of the present invention:

In step 2506, the count word is cleared. For example, when counter 1028receives a falling edge on input signal 2100, it clears count word 1074.

In step 2508, steps (a)–(c) are repeated n times for subsequentcorresponding cycles of the input signal, wherein n is equal to thenumber of bits of the control word. For example, as described above, foreach test or calibration cycle on input signal 2100, a successive bit ofcontrol word 1070 in SAR 1022 is adjusted based upon the value of countword 1074, until all bits of control word 1070 have been adjusted. Inalternative embodiments, a subset of the bits of control word 1070 areadjusted, instead of all bits.

In an embodiment, step 2508 may include the step where adjusting adifferent bit of the control word is adjusted each time that step (b)(2)is repeated, wherein the bit of the control word is adjusted accordingto at least one bit of the count word. For example, as described above,each bit of control word 1070 is adjusted according to the value of bit8 of count word 1074. For example, this may include the steps where thebit of the control word is set if the at least one bit of the count wordis equal to a first state, and the bit of the control word is reset ifthe at least one bit of the count word is equal to a second state. Inthe example provided above, a bit of control word 1070 is set if bit 8of count word is 1, and the bit of control word 1070 is reset if bit 8of count word 1074 is a 0. The present invention is applicable to one ormore of any of the bits of count word 1074 being used by SAR 1022 toadjust control word 1070.

FIG. 25C illustrate an additional step for flowchart 2500, according tofurther embodiments of the present invention:

In step 2510, each calibration cycle of the input signal followed with aseparation cycle on the input signal. For example, the separation cyclemay be separation cycle 2908 as shown in FIG. 29.

FIG. 25D shows a flowchart 2520 providing a algorithm for calibrating anoscillator frequency with an input signal, similar to that of FIGS.25A–C, according to embodiments of the present invention. FIGS. 25B–Cprovide steps according to further embodiments. Other structuralembodiments will be apparent to persons skilled in the relevant art(s)based on the following discussion. These steps are described in detailbelow.

In embodiment according to flowchart 2520, the oscillator frequency iscalibrated according to an input signal. For example, the oscillatorfrequency is the frequency of master clock signal 1062, and the inputsignal may be input signal 2100. Input signal 2100 includes calibrationor test waveforms of period t_(c).

Flowchart 2520 begins with step 2522. In step 2522, operation offlowchart 2520 begins. During step 2522, the contents of SAR 1022,control word 1070, are cleared, and the bit position of interest “m” ofSAR 1022 is set to the 0 bit position, which may be the MSB of SAR 1022,for example.

In step 2524, the value of bit position m in the contents of SAR 1022 isset equal to a “1” bit. For example, in the first iteration of flowchart2520, the bit position of m=0 in SAR 1022 is set equal to a “1” bit. Insubsequent iterations, subsequent bit positions will be set equal to a“1” bit in step 2524.

In step 2526, counter 1028 counts at its clock rate, which is thefrequency of master clock signal 1062 as shown in FIG. 21C, or may adivided frequency of master clock signal 1062 such as first clock signal1066 as shown in FIG. 21B. Counter 1028 counts for one cycle t_(c) ofthe input signal 2100, which is one cycle of a calibration or testwaveform as described above, to generate the value N on count word 1074.

In step 2528, the MSB of the counter word 1070, P₀, is checked todetermine whether it equals a “1” bit. If P₀ equals a “1” bit, thismeans that oscillator 1026 is counting too fast (i.e., f_(o)>f_(c)), andoperation proceeds to step 2532. If P₀ does not equal a “1” bit, thismeans that oscillator 1026 is counting at the proper rate, or too slow(i.e., f_(o)≦f_(c)), and operation proceeds to step 2530.

In step 2530, bit m of the contents of SAR 1022 is set to a “0” bit, andoperation proceeds to step 2532.

In step 2532, bit m is checked to determine whether the last bitposition of interest “m” of SAR 1022 has been processed, where the lastbit position in this example is the LSB of control word 1070. If thelast bit has been processed, operation proceeds to step 2536. If thelast bit has not been processed, operation proceeds to step 2534.

In step 2534, the bit position of interest “m” of SAR 1022 isincremented. For example, at the end of the first iteration of flowchart2520, bit position “m” is incremented from a “0” bit to the “1” bitposition. On the second iteration of flowchart 2520, bit position “m” isincremented from the “1” bit position to the “2” bit position of thecontents of SAR 1022. On the last iteration of 2520, bit position “m” isincremented to the MSB bit position of SAR 1022. In this manner, all bitpositions of SAR 1022 may be processed. Note that in alternativeembodiments, only a portion of the bit positions of SAR 1022 may beprocessed.

In step 2536, the calibration process shown in flowchart 2520 isfinished, and operation ends. Hence, the contents of SAR 1022 should beconfigured such that control word 1070 causes oscillator 1026 to outputthe desired oscillator frequency on master clock signal 1062.

4.4 Data Symbol Calibration

As discussed in section 1.2.1, reader network 104 transmits informationin the form of one or more symbols that are each selected from a symbolset. Tag 102 receives the transmitted symbols, and determines whatinformation the transmitted symbols represent. As shown in FIGS. 3–5,for example, a set of three symbol waveforms of varying duty cycles maybe used to represent three different logical values. The three logicalvalues that are represented by the waveforms of FIGS. 3–5 may be “0,”“1,” and “NULL,” for instance.

According to the present invention, the duration or length of timingintervals of waveforms that define the data symbols are set during acalibration routine. According to an embodiment, reader network 104transmits a series of pulse waveforms that are received by tag 102. Tag102 uses the received pulse waveforms to set boundaries for timingintervals that define data symbols. After tag 102 sets the data symbolstiming intervals, data waveforms subsequently received by tag 102 willbe compared to these timing intervals, to determine which logical valuesthe received data waveforms represent.

Note that in embodiments, a variety of characteristics of calibrationwaveforms received by a tag 102 from a reader network 104 may be used todefine data symbols during the calibration routine. For example, inembodiments, in addition to using a length or duration of a pulsewaveform to define data symbol timing intervals, amplitude, frequency,and phase of calibration waveforms transmitted by reader network 104 totags 102 may be used to define data symbols by tags 102.

FIG. 26A illustrates example waveforms that may be received by tag 102to calibrate data symbols, according to an embodiment of the presentinvention. FIG. 26A illustrates a first calibration waveform 2602, asecond calibration waveform 2604, and a third calibration waveform 2606.First calibration waveform 2602 corresponds to the timing parameter T0,described above. Second calibration waveform 2604 corresponds to thetiming parameter T1, described above. Third calibration waveform 2606corresponds to the timing parameter T2, described above.

In an embodiment, first, second, and third calibration waveforms 2602,2604, and 2606 are consecutively received by tag 102, and are used tocalibrate data symbols. First calibration waveform 2602 is firstreceived by tag 102. The duration or length of first calibrationwaveform 2602 is measured as the duration of the amount of time passingbetween the falling edge and rising edge of the pulse on firstcalibration waveform 2602. This is shown as T_(T0) in FIG. 26A (assumingthat T_(CS) is equal to zero). This length is stored by tag 102. Secondcalibration waveform 2604 is next received by tag 102. The duration orlength of the pulse on second calibration waveform 2604, shown as T_(T1)in FIG. 26A, is also stored by tag 102. Third calibration waveform 2606is lastly received by tag 102. The duration or length of the pulse onthird calibration waveform 2606, shown as T_(T2) in FIG. 26A, is alsostored by tag 102. After the three waveform pulse lengths are stored,they may be referred to, to determine the logical values for receiveddata symbols.

When the falling edge of a received data symbol pulse occurs (att_(cs)), the logical value for the data symbol may be determined byexamining the time period in which its trailing rising edge occurs. FIG.26A shows a first time period 2614, a second time period 2616, and athird time period 2618. First time period 2614 is a time period betweenT_(CS) and T_(T0). Second time period 2616 is a time period betweenT_(T0) and T_(T1). Third time period 2618 is a time period betweenT_(T1) and T_(T2). When the trailing rising edge of a data symbol pulseoccurs during first time period 2614, the data symbol will beinterpreted as a logical “0” value. When the trailing rising edge of adata symbol pulse occurs during second time period 2616, the data symbolwill be interpreted as a logical “1” value. When the trailing risingedge of a data symbol pulse occurs during third time period 2618, thedata symbol will be interpreted as a logical “NULL” value.

Hence, T_(T0) is a dividing line between logical “0” and logical “1”values. In an embodiment, T_(T0) may be equal to 4.5 μS, but may also beequal to shorter or longer amounts of time. T_(T1) is a dividing linebetween logical “1” and “NULL” values. In an embodiment, T_(T1) may beequal to 7.75 μS, but may also be equal to shorter or longer amounts oftime. Note that in an embodiment, T_(T2) indicates a time at which tag102 must stop transmitting data to a reader network 104. After T_(T2),tag 102 prepares for the falling edge of the next data symbol. In anembodiment, T_(T2) is equal to 11.5 μS, but may also be equal to shorteror longer amounts of time. For example, T_(T2) may be equal to a longertime period such as 24 μS, which allows reader network 104 to decreasetransmitted data rates in exchange for improved noise immunity.

Note that FIG. 26A also shows a first separator waveform portion 2608 offirst calibration waveform 2602, a second separator waveform portion2610 of second calibration waveform 2604, and a third separator waveformportion 2612 of third calibration waveform 2606. First, second, andthird separator waveform portions 2608, 2610, and 2612 are optional, andprovide time for tag 102 to store the received corresponding data symbolpulse, and to prepare for the next calibration/data pulse.

Examples of received data symbols are shown in FIGS. 3–5. As shown inFIG. 3, where the length T_(A) of a received data symbol is less thanT_(T0), the corresponding data symbol is interpreted as a logical “0”value. As shown in FIG. 4, where the length T_(B) of a received datasymbol is greater than T_(T0) and less than T_(T1), the correspondingdata symbol is interpreted as a logical “1” value. As shown in FIG. 5,where the length T_(C) of a received data symbol is greater than T_(T1)and less than T_(T2), the corresponding data symbol is interpreted as alogical “NULL” value.

FIG. 27 shows a data calibration and detection system 2700 in tag 102,according to an embodiment of the present invention. Data calibrationand detection system 2700 receives calibration signal pulses to performdata calibration, and also interprets received data symbols. Datacalibration and detection system 2700 includes counter 1028, a T0register 2702, a T1 register 2704, a T2 register 2706, a datacalibration logic 2708, and a data detection logic 2718. T0 register2702, T1 register 2704, T2 register 2706, data calibration logic 2708,and data detection logic 2718 may be included in state machine 1024,shown in FIG. 10, for example.

When performing data symbol calibration, counter 1028 measures lengthsof three calibration waveform pulses consecutively received on inputsignal 2100, and stores the pulse lengths in registers T0 register 2702,T1 register 2704, and T2 register 2706. Counter 1028 measures the lengthof a calibration waveform pulse according to the number of clock cyclesof first clock 1066 that occur between the falling and rising edges ofthe calibration waveform. The measured length is output on count word1074 and received by data calibration logic 2708. Data calibration logic2708 stores the received measured length in a respective register. Thelength of first calibration waveform 2602 is stored in T0 register 2702.The length of second calibration waveform 2604 is stored in T1 register2704. The length of third calibration waveform 2606 is stored in T2register 2706.

After data symbol calibration is completed, data symbols may be receivedon input signal 2100. When receiving a data symbol on input signal 2100,counter 1028 counts the length of the received data waveform accordingto first clock 1066. Counter 1028 begins counting when the received datawaveform transitions from high to low, and finishes counting when thereceived data waveform transitions from low to high. Counter 1028outputs count word 1074, which is received by data detection logic 2718.Data detection logic 2708 compares the length of the received datawaveform to the calibration waveform lengths stored in T0 register 2702,T1 register 2704, and T2 register 2706, to determine the logical valueof the received data. Data detection logic 2718 may determine thelogical values by direct comparison of the received data waveformlengths to the stored calibration waveform values, or in other ways.

For example, in an embodiment after data calibration has been performed,data symbols may now be transmitted to from reader network 104 to tag102. Data detection logic 2718 determines logical values for the datasymbols. Data detection logic 2718 receives count word 1074. After afalling edge on input signal 2100 count word 1074 is incremented upwardaccording to first clock signal 1066, for a duration of a received datasymbol. When count word 1074 equals the value stored in T0 register2702, data detection logic 2718 sets an internal T0 flag 2714. Whencount word 1074 equals the value stored in T1 register 2704, datadetection logic 2718 sets an internal T1 flag 2716. After receiving arising edge on input signal 2100, which indicates an end of the receiveddata symbol, the logical value for the received data symbol isdetermined by examining flags 2714 and 2716. If T0 flag 2714 is not set,the data symbol is interpreted as a logical “0” value. If T0 flag 2714is set, but T1 flag 2716 is not set, the data symbol is interpreted as alogical “1” value. If T0 and T1 flags 2714 and 2716 are both set, thedata symbol is interpreted as a logical “NULL” value. The interpretedlogical value for the received data symbol is output on interpreted datasignal 2710. Note that after the falling edge of a data symbol occurs,flags 2714 and 2716 are reset or initialized to be used to interpret thedata symbol being received.

FIG. 28A shows a flowchart 2800 providing steps for performing datacalibration, according to embodiments of the present invention. FIGS.28B–D provide steps according to further embodiments. The steps of FIGS.28A–D do not necessarily have to occur in the order shown, as will beapparent to persons skilled in the relevant art(s) based on theteachings herein. Additional structural embodiments for performing thesteps of FIG. 28A–D will be apparent to persons skilled in the relevantart(s) based on the following discussion. These steps are described indetail below.

Flowchart 2800 begins with step 2802. In step 2802, a first calibrationpulse is received on an input signal. For example, the first calibrationpulse is the pulse of first calibration waveform 2602, which is receivedon input signal 2100. The first calibration pulse may be received atcounter 1028, as shown in FIG. 27.

In step 2804, a length of the first calibration pulse is stored. Forexample, counter 1028 determines the length of the pulse of firstcalibration waveform 2602, and outputs the length of the pulse on countword 1074. T0 register 2702 receives count word 1074, and stores thelength of the pulse of first calibration waveform 2602.

In step 2806, a second calibration pulse is received on the inputsignal. For example, the second calibration pulse is the pulse of secondcalibration waveform 2604, which is received on input signal 2100. Thesecond calibration pulse may be received at counter 1028.

In step 2808, a length of the second calibration pulse is stored. Forexample, counter 1028 determines the length of the pulse of secondcalibration waveform 2604, and outputs the length of the pulse on countword 1074. T1 register 2704 receives count word 1074, and stores thelength of the pulse of second calibration waveform 2604.

FIG. 28B illustrates additional steps for flowchart 2800 of FIG. 28A,according to further embodiments of the present invention. In theembodiment described in reference to FIG. 28A, the data symbolcalibration procedure receives and stores two data calibration pulses.FIG. 28B describes the detection of a received data symbol using thereceived and stored data calibration pulses.

In step 2810, a data symbol having a pulse portion is received on theinput signal, wherein the pulse portion has a third length. For example,the data symbol may be one of the received data symbols shown in FIGS.3–5. The pulse portion is the data symbol portion between falling andrising edges of the pulse of the respective waveform of FIGS. 3–5, suchas T_(A), T_(B), and T_(C). Hence, the length of the pulse portion ofthe data symbols shown in FIGS. 3–5 may be T_(A), T_(B), and T_(C),respectively.

In step 2812, a first flag is set if the length of the pulse portion isgreater than or equal to the stored length of the first calibrationpulse. For example, the first flag may be T0 flag 2714 of datacalibration logic 2708, as shown in FIG. 27. Data calibration logic 2708performs a comparison of the incrementing value of count word 1074 tothe contents of T0 register 2702, and sets T0 flag 2714 if they becomeequal. For illustrative purposes, for steps 2812 through 2820, T_(T0) isassumed to be equal to 4.5 μS and T_(T1) is assumed to be equal to 7.75μS. In this example, T0 flag 2714 would become set during step 2812 whenreceiving the data symbols shown in FIGS. 4 and 5, which have respectivelengths of 6 μS and 9.5 μS.

In step 2814, a second flag is set if the third length is greater thanor equal to the stored length of the second calibration pulse. Forexample, the second flag may be T1 flag 2716 of data calibration logic2708, as shown in FIG. 27. Data calibration logic 2708 performs acomparison of the incrementing value of count word 1074 to the contentsof T1 register 2704, and sets T1 flag 2716 if they become equal. In thecurrent example, T1 flag 2716 would become set when receiving the datasymbol shown in FIG. 5, which has a length of 9.5 μS.

In step 2816, the third pulse is determined to be a first logical valueif the first flag is not set during step 2812. In the current example,the data symbol shown in FIG. 3 is determined to be a first logicalvalue because T0 flag 2714 was not set during step 2812.

In step 2818, the third pulse is determined to be a second logical valueif the first flag is set and the second flag is not set. In the currentexample, the data symbol shown in FIG. 4 is determined to be a secondlogical value because T0 flag 2714 was set during step 2812, and T1 flag2716 was not set during step 2814.

In step 2820, the third pulse is determined to be a third logical valueif the first flag is set and the second flag is set. In the currentexample, the data symbol shown in FIG. 5 is determined to be a thirdlogical value because T0 flag 2714 was set during step 2812, and T1 flag2716 was set during step 2814.

In step 2822, the first logical value is defined as a logical “0” bit.In embodiments, the first logical value may alternatively be defined asa logical “1” bit, a “NULL” bit, or other logical value.

In step 2824, the second logical value is defined as a logical “1” bit.In embodiments, the second logical value may alternatively be defined asa logical “0” bit, a “NULL” bit, or other logical value.

In step 2826, the third logical value is defined as a logical “NULL”bit. In embodiments, the third logical value may alternatively bedefined as a logical “0” bit, a logical “1” bit, or other logical value.

FIG. 28C illustrates additional steps for flowchart 2800 of FIG. 28A,according to further embodiments of the present invention. Similarly tothe embodiment described in reference to FIG. 28B, the data symbolcalibration procedure of FIG. 28C only requires two data calibrationpulses to be received and stored:

In step 2828, a data symbol having a pulse portion is received on theinput signal, wherein the pulse portion has a length. For example, thedata symbol may be one of the received data symbols shown in FIGS. 3–5.The pulse portion is the data symbol portion between falling and risingedges of the pulse of the respective waveform of FIGS. 3–5, such asT_(A), T_(B), and T_(C). Hence, the length of the pulse portion of thedata symbols shown in FIGS. 3–5 may be T_(A), T_(B), and T_(C),respectively.

In step 2830, the data symbol is determined to be a first logical valueif the length of the pulse portion is less than the stored length of thefirst calibration pulse. For example, data calibration logic 2708compares the length of the pulse portion to the value stored in T0register 2702. If the length of the pulse portion is less than the valuestored in T0 register 2702, the data symbol is determined to the firstlogical value. For illustrative purposes, for steps 2830 through 2834,T_(T0) is assumed to be equal to 4.5 μS and T_(T1) is assumed to beequal to 7.75 μS. In this example, when the data symbol is the datasymbol shown in FIG. 3, which has a length of 3 μS, the data symbolwould be determined to be the first logical value. This is because thelength of the pulse shown in FIG. 3 is less than the length of the pulseof first calibration waveform 2602 (i.e., 4.5 μS), which is stored in T0register 2702.

In step 2832, the data symbol is determined to be a second logical valueif the length of the pulse portion is greater than or equal to thestored length of the first pulse and less than the stored length of thesecond calibration pulse. For example, data calibration logic 2708compares the length of the pulse portion to the value stored in T0register 2702 and the value stored in T1 register 2704. If the length ofthe pulse portion is greater than or equal to the value stored in T0register 2702, and less than the value stored in T1 register 2704, thedata symbol is determined to the second logical value. In this example,when the data symbol is the data symbol shown in FIG. 4, which has alength of 6 μS, the third pulse would be determined to be the secondlogical value. This is because the length of the pulse shown in FIG. 4is greater than the length of the pulse of first calibration waveform2602 (i.e., 4.5 μS), which is stored in T0 register 2702, and less thanthe length of the pulse of second calibration waveform 2604 (i.e., 7.75μS), which is stored in T1 register 2704.

In step 2834, the data symbol is determined to be a third logical valueif the length of the pulse portion is greater than the stored length ofthe second pulse. For example, data calibration logic 2708 compares thelength of the pulse portion to the value stored in T1 register 2704. Ifthe length of the pulse portion is greater than or equal to the valuestored in T1 register 2704, the data symbol is determined to the thirdlogical value. In this example, when the data symbol is the data symbolshown in FIG. 5, which has a length of 9.5 μS, the data symbol would bedetermined to be the third logical value. This is because the length ofthe pulse shown in FIG. 5 is greater than the length of the pulse ofsecond calibration waveform 2604 (i.e., 7.75 μS), which is stored in T1register 2704.

FIG. 28D illustrates additional steps for flowchart 2800, according tofurther embodiments of the present invention. The data symbolcalibration procedure of FIG. 28D receives and stores three datacalibration pulses:

In step 2836, a third calibration pulse is received on the input signal.For example, the third calibration pulse is the pulse of thirdcalibration waveform 2606, which is received on input signal 2100. Thethird calibration pulse may be received at counter 1028, as shown inFIG. 27.

In step 2838, a length of the third calibration pulse is stored. Forexample, counter 1028 determines the length of the pulse of thirdcalibration waveform 2606, and outputs the length of the pulse on countword 1074. T2 register 2706 receives count word 1074, and stores thelength of the pulse of third calibration waveform. The value stored inT2 register 2706 may have a variety of uses. For example, in anembodiment, T_(T2) indicates a time at which tag 102 must stoptransmitting data to a reader network 104. After T_(T2), tag 102prepares for the falling edge of the next data symbol.

In embodiments, after performing steps 2836 and 2838 shown in FIG. 28D,data symbols may be received and evaluated to determine their logicalvalues, as is described above.

FIGS. 28E–F show flowcharts that provide more detailed exemplary stepsfor performing data detection as described in FIGS. 28B–C, according toembodiments of the present invention. The steps of FIGS. 28E–F do notnecessarily have to occur in the order shown, as will be apparent topersons skilled in the relevant art(s) based on the teachings herein.Additional structural embodiments for performing the steps of FIG. 28E–Fwill be apparent to persons skilled in the relevant art(s) based on thefollowing discussion. In particular, the embodiment shown in FIG. 28Eshows more detail of flowchart 2800 as shown in FIG. 28B, while theembodiment shown in FIG. 28F shows more detail of flowchart 2800 asshown in FIG. 28C. A flowchart shown in FIG. 28E is described asfollows:

In step 2850, a T0 flag and a T1 flag are reset. For example, the T0flag is T0 flag 2714 and the T1 flag is T0 flag 2716.

In step 2852, a pulse, such as a data symbol pulse portion, is receivedfrom a reader in reader network 104.

In step 2854, the length of the pulse is counted by a counter. Forexample, counter 1028 counts the length of the pulse portion of a datasymbol according to master clock signal 1062, first clock signal 1066,or second clock signal 1064.

In step 2856, the counter contents are stored in T_(count). T_(count)represents the length of the received data symbol pulse portion.

In step 2858, T_(count) is compared to the stored value for T0. IfT_(count) is greater than or equal to T0, operation proceeds to step2860, where the T0 flag is set, and operation proceeds to step 2862. IfT_(count) is not greater than or equal to T0, operation proceeds to step2866.

In step 2862, T_(count) is compared to the stored value for T1. IfT_(count) is greater than or equal to T1, operation proceeds to step2864, where the T1 flag is set, and operation proceeds to step 2866. IfT_(count) is not greater than or equal to T1, operation proceeds to step2866.

In step 2866, if the T1 flag is set, operation proceeds to step 2868,where the received signal is determined to be a “NULL” symbol. If the T1flag is not set, operation proceeds to step 2870.

In step 2870, if the T0 flag is set, operation proceeds to step 2872,where the received signal is determined to be a “1” symbol. If the T0flag is not set, operation proceeds to step 2874.

In step 2874, the received signal is determined to be a “0” symbol.

Operation proceeds from steps 2868, 2872, and 2874 to step 2876. In step2876, the determined symbol is returned to logic in tag 102 for anyapplicable use.

FIG. 28F shows a flowchart similar to that shown in FIG. 28E, wheresteps 2860, 2864, 2866, and 2870 are not necessary.

Note that the above described embodiments refer to the data symbols anddata calibration pulses as being “negative-going” pulses, which have afalling edge followed by a rising edge. However, the present inventionis also applicable to the data symbols and data calibration pulses being“positive-going” pulses, having a rising edge followed by a fallingedge. The present invention is also applicable to any combination ofnegative-going and positive-going pulse types for signals, including thedata symbols and data calibration pulse signals.

The embodiments above describe the transmission of two and threecalibration pulses to remotely program one or more tags 102 to interprettwo or more data symbol types (e.g., logical “0,” “1,” and “NULL” datasymbols). Furthermore, the present invention is applicable to the use ofadditional calibration pulses to remotely program one or more tags 102to interpret any number of additional data symbol types. For example,calibration pulses may be transmitted to define the timing boundaries inone or more tags 102 for data symbols that represent multiple bits, suchas “00,” “01,” “10” “11,” “001,” and for further data symbols.

5. Front End Embodiments of the Present Invention

FIG. 30 illustrates an analog front-end 3000 of an exemplary RF tagdescribed herein. The analog front 3000 includes an antenna 3002, an RFpad 3004, an electrostatic device (ESD) protection circuit 3006, a powercharge pump 3008, a data recovery circuit 3010, and a backscatter switch3016. The antenna 3002 receives a RF signal 3003 that is transmitted bythe tag reader, and provides the RF signal 3003 to the RF pad 3004. TheRF signal 3003 can be modulated to carry information, e.g. pollingquery, from the reader to the tag. As will be discussed further herein,the energy in the RF signal 3003 can also be harvested to power theanalog front 3000 and the rest of the tag.

The ESD protection circuit 3006, the power charge pump 3008, and thedata recovery circuit 3010 are connected to the RF pad 3004, and arearranged in-parallel with each other at node 3005. This enables thementioned circuits to simultaneously process the RF signal 3003 for bothdata recovery and for power generation. The ESD protection circuit 3006provides over-voltage protection for the analog front 3000, such as fromstatic discharge. The ESD protection circuit 3006 also providesprotection from RF signals 3003 that could damage the analog front 3000.For example, this can occur when a tag is very close, or adjacent to, atransmitting reader. The power charge pump 3008 converts the RF signal3003 to a DC voltage V_(DD) at an output terminal 3009. Morespecifically, the power charge pump 3008 rectifies the RF signal 3003,increases the voltage amplitude, and generates an output voltage V_(DD).The DC voltage V_(DD) is sufficiently stable so that it can be used as avoltage supply for the analog front 3000, and also for the rest of thetag. The data recovery circuit 3010 process the RF signal 3003 inparallel with the power charge pump 3008 and generates a digital outputsignal at the output terminal 3012. The back scatter switch 3016receives a modulated data signal at a terminal 3017, and gates theimpedance of the antenna 3002 to transmit the modulated data signal backthe reader. More specifically, the backscatter switch 3016 shorts theinput impedance seen at the RF pad 3004, so that the RF signal 3003 isreflected back to the reader in accordance with the modulated datasignal that is received at the terminal 3017.

The analog front-end 3000 in FIG. 30 can be related back to the tag 1001that is shown in FIG. 10. More specifically, in one embodiment, thereceiver 1030 is the data recovery circuit 3010. The modulator 1034 isthe back scatter switch 3016. The charge pump 1032 is the main chargepump and RF clamp 3008. The RF pad 1004 is the RF pad 3004. The antenna1010 is the antenna and Z-match 3002.

5.1 Power Charge Pump

FIG. 31 illustrates a power charge pump 3100 that is one embodiment ofthe power charge pump 3008 of the analog front-end 3000. The powercharge pump 3100 rectifies the RF signal 3003 received at an inputterminal 3101, increases the voltage amplitude, and generates the outputDC voltage V_(DD) at an output terminal 3114. The output voltage V_(DD)is sufficiently stable that it can be used as a voltage supply for therest of the tag, including the data recovery circuit 3010. Morespecifically, once a steady state voltage is reached, further increasesin the power level of the RF signal 3003 produce smaller increases theoutput DC voltage V_(DD). This occurs because the efficiency of thecharge pump 3100 is designed to intentionally decrease once the RFsignal 3003 reaches a threshold power level.

The charge pump 3100 includes multiple stages 3102 a–d. Any number ofstages 3102 could be utilized, and four stages are shown in FIG. 31 forconvenience of discussion only. Each stage 3102 includes a diode 3104, adiode 3108, a diode 3110, and a capacitor 3106. The capacitor 3106 ineach stage is connected to a central node 3107 and to the input terminal3101 so that each stage 3102 simultaneously receives the input signal3003. The diodes 3104 and 3108 in each stage are connected betweenground and the central node 3107. More specifically, the anode of thediode 3104 is connected to the node 3111 in the prior stage 3102 (expectfor the first stage 3102 a where the anode of diode 3104 a is connecteddirectly to ground), and the cathode of the diode 3104 is connected tothe central node 3107. The diode 3110 in each stage is connected betweenthe node 3107 and a second node 3111, which connects to the following oradjacent stage 3102. More specifically, the anode of the diode 3110 isconnected to the central node 3107, and the cathode of the diode 3110 isconnected to the second node 3111.

FIGS. 32A–32C illustrate the desired current voltage curves (i.e. IVcurves) for the respective diodes 3110, 3104, and 3108. All the diodes3110, 3104, and 3108 have respective forward threshold voltage V_(TH0),V_(TH1), and V_(TH2), where the current begins to flow in the forwarddirection once the forward threshold voltage is reached. Comparing FIGS.32A–32C, V_(TH1)<V_(TH0)<V_(TH2). Therefore, the diodes 3104 shouldconduct before the diodes 3110 in a low power RF environment. As will bediscussed further herein, the diodes 3108 are intended to bleed chargefrom the charge pump 3100 when the RF power level reaches a thresholdlevel. Hence, the V_(TH2) is set higher than V_(TH0) or V_(TH1), so thecharge is not bled off too soon. Additionally, the diodes 3110, 3104,and 3108 have respective breakdown voltages V_(B1), V_(B2), and V_(B3),where charge begins to flow in the reverse direction once the reversebreakdown voltage is reached.

As will be discussed further herein, the diodes 3104, 3108, and 3110 canbe implemented using MOSFET equivalent diode structures that approximatethe operation of a diode. In one embodiment, the traditional MOSFETdevice for the diodes 3110 and 3104 is modified to increase theconductivity of the diode, and to cause the diode to conduct at a lowerRF signal level.

The operation of the charge pump 3100 is as follows. The RF signal 3003is simultaneously applied to each stage 3102 through the capacitors3106. During a positive cycle of the RF signal 3003, the capacitor 3106in each stage transfers charge to the node 3107. The diode 3110 isforward biased by the charge on the node 3107, causing the diode 3110 toconduct and transfer the charge from the node 3107 to the node 3111. Thecharge on the node 3111 is stored on the capacitor 3112 until the nextpositive RF cycle. The diode 3104 is reversed biased during the positivecycle and therefore do not conduct any charge. The diode 3108 is forwardbiased, but does not conduct any charge unless the charge on node 3107exceeds its increased threshold voltage. During the negative cycle ofthe RF signal 3003, the diode 3104 is forward biased and conducts chargefrom the node 3111 in one stage 3102 to the node 3107 in an adjacentstage 3102 (except for 3104 a which also transfers charge from ground tonode 3107 a). Charge is also transferred from ground to the node 3111through the capacitor 3112. The diodes 3110 and 3108 are reversed biasedand do not conduct any charge. During the next positive cycle, the diode3110 is again forward biased, moving charge from node 3107 to the node3111 within each stage. The charge that is moved from the node 3107 tothe node 3111 includes both the charge accumulated on the node 3107during the positive cycle, but also the charge the accumulated on thenode 3107 from the negative cycle. Over multiple cycles of the RF signal3003, charge accumulates and increases as moves it through the stages3102 a–3102 d, and the corresponding voltage is added in-series at thecapacitors 3112. The accumulated charge at the output node 3112 e isconverted to a DC voltage V_(DD) by the final capacitor 3112 e. The DCvoltage V_(DD) can be tapped and used as voltage supply at the outputterminal 3114.

As stated above, the charge steadily increases as it moves through thestages 3102 a to 3102 d to the output node 3114 because of theadditional charge added by each stage 3102 during each full cycle of theRF signal 3003. Therefore, the charge at the node 3114 increases withthe number of stages 3102, and causes the DC voltage V_(DD) at the node3114 to have a larger amplitude than the amplitude of the RF signal3003. Theoretically, the DC voltage V_(DD) would grow at infinitum withthe number of stages 3102. However, parasitic resistance andcapacitances of real circuit elements begin to become significant asmore and more stages are added, reducing the overall efficiency of thecharge pump. Furthermore, the overall circuit size also increases withthe number of stages. Experiments have shown that four stages are anefficient design, but any number of stages could be utilized based onthe specific circuit application.

It is desirable for the output voltage V_(DD) to be relatively stablevalue since it is being used as a voltage supply for the rest of thetag. The charge that is accumulated at each stage 3102 is directlyproportional to the power level of the RF signal 3003. In other words,the larger the amplitude of the RF signal 3003, the more charge istransferred to the nodes 3107 and 3111 in each stage 3102. Therefore,without regulation, large RF power levels could generate a DC voltageV_(DD) that is sufficiently large to damage the components of the tag.However, the diodes 3108 prevent this from happening by conducting whenthe amplitude of the RF signal 3003 becomes so large that the voltage atthe nodes 3107 exceeds the threshold voltage V_(TH2) of the diodes 3108.The threshold voltage of the diodes is set so that these diodes conductfor a given RF power level. When the diodes 3108 conduct, then charge isremoved or “bled” from the nodes 3107 to ground through the diodes 3108.The charge that is bled-off to ground does not make it to the outputnode 3114, and therefore does not increase the DC voltage at the outputnode 3114. Furthermore, once the diodes 3108 begin to conduct, aresistive RF ground is placed at the nodes 3107, causing an impedancemismatch at the input terminal 3101. The impedance mismatch de-tunes theantenna 3002 causing it to reflect some of the RF signal 3003 away. Theresult is that the overall efficiency of the charge pump 3100 begins todecrease once the diodes 3108 begin to conduct, regulating the outputvoltage V_(DD) to a pre-determined voltage.

FIGS. 33A and 33B illustrate the DC output voltage and charge pumpefficiency verses the RF input power when using the diodes to limit theoutput voltage of the charge pump 3100. FIG. 33B illustrates theefficiency vs. RF input power, and illustrates that the efficiency peaksat an RF power level 3302, and then rolls off for RF power that isgreater than this. FIG. 33A illustrates the DC output voltage verses RFinput power, given the efficiency curves in FIG. 33B. As shown in FIG.33A, the output voltage rises non-linearly with the RF input power untilthe efficiency peak at 3302. At which point, the DC output voltage onlyincreases at a much slower linear rate.

FIG. 34 further illustrates one embodiment 3400 for the charge pump3100, where the diodes in each stage 3102 are replaced with metal oxidefield effect transistors (MOSFET) that are configured as diodeequivalents devices. For example, the diodes 3104 in each stage 3102 areconfigured as MOSFET diodes 3402, the diodes 3108 are configured assingle or stacked MOSFET devices 3404, and the diodes 3110 areconfigured as MOSFET diodes 3406. The terminals of the MOSFETs diodes3402 and 3406 are connected to produce diode equivalent circuits.Additionally, the terminals of the mentioned MOSFET diodes 3402 and 3406are connected so as effectively reduce the threshold voltage of theMOSFET diodes. This improves the operation of the charge pump 3100 in alow power environment, such as when the tag is far away from the reader.

The threshold voltage for the MOSFET diodes 3402 and 3406 is reduced byconnecting the body terminal, the source terminal, and gate terminal alltogether for each of the MOSFETs diodes. Whereas, in a conventionalMOSFET diode configuration, the source and body terminals are connectedtogether, but the gate terminal is connect to the drain terminal tocreate the two terminal diode circuit. By connecting the gate terminalwith the source and drain terminals, the characteristics of the MOSFETdiode are modified so that the threshold voltage of the MOSFET diode islowered, which causes the MOSFET diode to conduct in a lower powerenvironment than it normally would, and the MOSFET diode doesn't conductin the reverse direction. As a result, the charge pump 3100 is able togenerate a useful DC voltage at the output terminal 3114, even when thetag is located far away from the reader. The modified MOSFET diodestructure is compared with a conventional diode structure below.

The terminals of the MOSFETs 3404 are connected to so the MOSFETs 3404operate as regular MOSFET devices, as opposed to MOSFET diodes 3402 and3406. More specifically, in the MOSFET devices 3404, the respective gateterminals are not connected to the body and source terminals, as in theMOSFETs diodes 3402 and 3406. The threshold of the devices 3404 isincreased by either a fabrication process adjustment, or by stacking thedevices as shown in FIG. 34B. However, in this configuration, the MOSFETdevices 3404 conduct in the reverse direction (i.e. negative biasvoltage), so the MOSFET device 3404 does not operate as a true diode.Further details regarding the IV curves for the MOSFETs diodes 3402 and3404, and the MOSFET devices 3406 will be discussed herein.

FIGS. 35A and 35B illustrate an unbiased MOSFET 3500 having a gateterminal, a drain terminal, a body terminal, and a source terminal. FIG.35B illustrates a cross-sectional view a conventional MOSFET. Referringto FIG. 35B, an underlying substrate 3510 has a lightly p-doped well3512. The p-doped well 3512 has two N+ regions 3502 and 3506, a P+region 3508, and a gate oxide 3504 that is disposed on top of thep-doped well 3512. The gate terminal is connected to a conductive region3503, such as aluminum, that is disposed on top of the gate oxide layer3304. The drain and source terminals are connected to respective N+layer 3502 and 3506, and the body terminal is connected to the P+ region3508. It is noted that the drain and source designations areinterchangeable, and these labels are only utilized for convenience ofdiscussion. The source and drain of the MOSFET 3500 form N⁺/P diodeswith the p-well 3512, where the depletion region extends mostly in theP⁻ well 3512, as is shown in the FIG. 35B.

FIGS. 36A–36C illustrate a MOSFET 3600 biased as a conventional loaddevice. Referring to FIG. 36A, the body and source of the MOSFET device3600 are both connected to ground. The gate and drain are connectedtogether, and are either normally connected to +V. FIG. 36B illustratesthe conventionally biased MOSFET 3600 with a +V bias voltage on thedrain and gate terminals. The +V bias voltage draws carriers toward thegate oxide 3504 to form a conductive inversion region 3602 that carriescurrent from the drain 3502 to the source 3506. A depletion region 3604is formed by the vacated carriers that form the conductive channel 3602.The depletion region is generally devoid of carriers since they are usedto form the conductive channel 3602. FIG. 36C illustrates theconventionally biased MOSFET diode 3600 with a −V bias voltage on thedrain and the gate terminals. The −V bias voltage drives away carriersfrom the underneath the gate region 3504, so that no channel is formedand no current flows from the drain to the source terminals. However, areverse current does flow from the body terminal to the drain terminalsbecause the P doped well 3512 and the N+ region 3502 form a forwardbiased PN junction 3606. The reverse current is significant for DC andlow frequency voltage biases, preventing its use as an effective diode.But, as the frequency increases, the carrier response time in thedepletion region starts to limit the reverse current and the device actsas a leaky diode.

FIGS. 37A–C illustrates a MOSFET diode 3700 biased according to thepresent invention so as to lower the threshold voltage of MOSFET diodeconfiguration and to prevent reverse bias conduction. Herein, thisconfiguration is referred to as the “back-bias” effect. Referring toFIG. 37A, the drain, body, and gate terminals of the MOSFET 3700 areconnected together and connected to the bias voltages +V and −V thatturn the diode on or turn the diode off. The source terminal of theMOSFET 3700 is connected to ground. The terminal connections of theMOSFET 3700 are different from that of MOSFET 3600 because the bodyterminal in the MOSFET 3700 is biased to the same voltage potential asthe gate terminal. Whereas, in the conventional MOSFET diode 3600, thebody terminal is not connected to the gate terminal, but is connected tothe source terminal. By biasing the body to the same potential as thegate, the characteristics of the MOSFET 3700 are changed so as to lowerthe threshold voltage of the diode, as will be discussed with referenceto FIGS. 37B and 37C.

Referring to FIG. 37B, a +V bias voltage on the gate 3504 draws carrierstoward the gate oxide 3504 to form a conducting inversion channel 3702that carries current from the drain 3502 to the source 3506. A depletionregion 3706 is formed by the vacated carriers that form the conductivechannel 3702. The depletion region 3706 tapers off near the source 3506because the body terminal 3508 and the p-doped well 3512 and the gateare biased to V+. The result of the reduced depletion region 3706 nearthe source 3506 is that the threshold voltage is reduced for the MOSFETdiode 3700 through the “back bias effect”. The “back bias effect” inMOSFETs is further in the “Physics of Semiconductor Devices”, S. M. Sze,published by J. Wiley and Sons, 1981, pages 438–445, which isincorporated herein by reference. Therefore, the MOSFET diode 3700begins to conduct for bias voltages +V, which equates to lower power RFsignals in the tag environment. In addition to the conductive channel3702, the +V bias voltage on the body terminal produces a forward biasedjunction 3704 because the p-doped well 3512 is forward biased relativeto the source 3506, causing forward bias diode current to flow from thebody terminal to the source terminal. This additional current is onlyresponsive to low frequency or DC bias voltages, and responds poorly tohigh frequency RF signals because the diode junction carriers do notrespond quickly enough to follow these higher frequencies.

FIG. 37C illustrates the MOSFET diode 3700 with a −V bias voltageapplied to the body, drain, and gate terminals. The −V bias voltagedrives away carriers from the underneath the gate region 3504 andgenerates an enlarged depletion region 3708. The enlarged depletionregion 3708 prevents current from flowing between the drain terminal3502 to the source terminal 3506. It is noticed that there is no reversecurrent flow between the body 3508 to the source 3506 because there isno forward biased junction between the P+body 3508 and the N+ source3506. In contrast, the conventional MOSFET device 3600 suffers a reversecurrent caused by the forward biased diode junction 3606. This reversecurrent is undesirable because it prevents true diode behavior.

FIGS. 38A–C compare the IV curve for the MOSFET diode 3700 with the IVcurve of the conventional MOSFET device 3600, and with the IV curve of aSchottky diode. More specifically, FIG. 38A illustrates an IV curve 3802for the MOSFET diode 3700. FIG. 38B illustrates an IV curve 3804 for theMOSFET device 3600. Finally, FIG. 38C illustrates an IV curve 3806 for aSchottky diode. The IV curve 3806 for the Schottky diode has a fixedslope that is much steeper than the slope of the conventional MOSFETdevice 3600. The steeper slope of the Schottky is desirable as itequates to low-power conduction, but the Schottky performance istypically not attainable in a standard MOSFET process.

Comparing FIG. 38A to FIG. 38B, the IV curve 3802 for the MOSFET diode3700 is steeper than the IV curve 3804 of the MOSFET device 3600.Furthermore, the threshold voltage (V_(TH)) for the MOSFET diode 3700 islower than the corresponding threshold voltage (V_(TH)) for the MOSFETdevice 3600. These improvements are caused by the back bias effect andthe forward biased diode configuration. The back bias reduces thethreshold voltage for the MOSFET diode 3700 by the square root its valueso the threshold continues decrease as the back bias is increased. Theoverall result is that the IV curve 3802 for the MOSFET diode 3700 issteeper the corresponding IV curve 3804 for the conventional MOSFETdevice 3600, but not as steep as the Schottky diode curve 3806.Accordingly, the MOSFET diode curve 3700 enables the charge pump 3400 tooperate at a much lower RF power levels than that which can be achievedwith a conventional MOSFET diode 3600 because the diode 3700 is moreconductive.

The back bias is further described by the following equation:I _(D) ≅[V−(V _(TH) −∂√{square root over (V)})]²

where:

-   -   ∂is a process constant with a value less than 1;    -   V is the bias voltage; and    -   V_(TH) is the threshold voltage.

As seen by the equation above, the back bias configuration effectivelyreduces the threshold voltage V_(TH), by the factor ∂√{square root over(V)}, where V is the bias voltage. Therefore, the effective thresholdvoltage decreases as the bias voltage increases, which causes the IVcurve to steepen and increases current conduction.

In the charge pump 3400, the bias voltage for the MOSFET diode 3700 isthe RF input signal 3003. Therefore, the bias voltage +V/−V varies withthe sinusoidal cycle of the RF signal 3003 and with the amplitude (i.e.power level) of the RF 3003, which varies with relative location of thetag from the reader in the tag environment. Because of its dynamicthreshold voltage, the leakage characteristics of the MOSFET diode 3700are not degenerated in the reverse direction as they would be if thethreshold voltage was permanently lowered by adjusting the semiconductordoping levels. For instance, FIG. 39 illustrates the effect of loweringthe threshold voltage for the conventional MOSFET 3600 by adjusting thedoping levels. The IV curve 3902 is associated with a first dopinglevel, and the IV curve 3904 is associated with a second doping level.As shown, the IV curve 3902 has a lower threshold voltage than the IVcurve 3904. However, the IV curve 3902 also has more leakage current inthe reverse (i.e. negative voltage) direction than the IV curve 3904.Absent output limiting, this leakage current is generally undesirablebecause it reduces the efficiency of the charge pump 3400 since chargeis leaked to ground. Referring back to FIG. 38, the MOSFET diode 3700does not suffer from increased leakage current because the thresholdvoltage is dynamic as described above.

FIG. 40 illustrates a charge pump 4000 according to another embodimentof the present invention. The charge pump 4000 is similar to the chargepump 3400 except that at least one of the MOSFET diodes in each stage3102 is forward biased. More specifically, the first series diode 3402in each stage 3102 is forward biased with the output of the adjacentsecond series diode 3404. For instance, the gate of MOSFET 3402 a isbiased with the voltage on node 3111 a, which is the output of thesecond diode 3404 a. Node 3111 a can be described as the output node ofthe first stage 3102 a, or equivalently, node 3111 a can be described asthe input node of the second stage 3102 b. Similarly, the gate of theMOSFET 3402 b is biased with the voltage on the node 3111 b, which isthe output of the MOSFET diode 3404 b, and so on for the remainingMOSFET diodes. To summarize, the gate of the n^(th) series diode isforward biased with the output of the (n+1)^(th) series diode, forn=odd. Stated another way, every other series diode has its gateforwarded biased with the output of the adjacent series diode that has ahigher output voltage.

Still referring to FIG. 40, the charge and voltage in the charge pump4000 accumulates along with the number of stages 3102 to produce theoutput voltage V_(DD) at the output node 3111 d. Therefore, using fourstages 3102, the voltage should increase approximately by ¼ V_(DD) atthe output of each stage 3102. Therefore, by connecting the gate voltageof MOSFET 3402 a with the output of the MOSFET 3404 a, the gate voltageon the MOSFET 3402 a is boosted by ¼ V_(DD) greater than what it wouldotherwise be if it was connected to ground, as in charge pump 3400. Asimilar boost in gate voltage occurs for the MOSFET diodes 3404 b, 3404c, and 3404 d.

The effect of forward biasing the gate voltage on the series MOSFETdiodes 3402 is to shift the operating point of these the diodes byV_(DD)/4 on their respective IV curves. FIGS. 41A and 41B illustrate anexemplary IV curve 4100 to illustrate this point. If the diode 3402 isoperating over a bias voltage range 4102 (FIG. 41A) for a give RF signallevel, then the forward biased gate will effectively move the operatingbias voltage to an effective bias voltage 4104 (FIG. 41B) on the curve4100. For example, the range 4104 can be shifted-up by V_(DD)/4, causingmore conduction over range 4104 compared to range 4102. Therefore, aforward biased diode 3402 will be more conductive then it otherwisewould be. Accordingly, the charge pump 4000 will operate with a lower RFinput signal 3003 than it otherwise would. However, the diodes will bemore leaky in the reverse direction, since during part of the negativevoltage cycle of the sine wave, the diodes will still have positive gatebias, allowing current to flow. However, this can be used to limit theadvantage of the charge to prevent over-voltage.

As discussed above, the charge pump 4000 is configured so that everyother series diode is forwarded biased with the adjacent series diode.However, the invention is not limited to this configuration, as anynumber of series diodes (3402 or 3404) could have their gates forwardedbiased with the output of another series diode (3402 or 3404). Forinstance, every series diode (3402 and 3404) could have its gateforwarded biased, if so desired. To generalize this embodiment, assumingthe series diodes are numbered from 1-to-n for convenience, then thegate of the 1^(th) diode can be forwarded biased with the output of them^(th) diode, assuming that 1<m≦n. These bias variations can change theperformance of the charge pump.

To summarize, the charge pump 4000 has at least two features thatimprove the efficiency of the charge pump operation. First, the gate ofeach MOSFET diode is connected to the same voltage potential as the bodyand the source or drain of the MOSFET diode. By connecting using thisconfiguration, the depletion layer of the MOSFET diode is modified so asto steepen the IV curve of each MOSFET diode, which improves theconductivity of the MOSFET diode. Second, the gate of at least oneMOSFET diode is forward biased with the output of an adjacent MOSFET,thereby raising the gate voltage and improving the conductivity of theforward-biased MOSFET diode.

5.2 Data Recovery

FIG. 42 illustrates a data recovery circuit 4200 that is one embodimentof the data recovery circuit 3010 in FIG. 30. The data recovery circuit4200 processes the RF signal 3003 in parallel with the power chargepump, and demodulates any information that is carried by the RF signal3003 to generate a digital output signal 4220 at the output terminal4218. The data recovery circuit 3010 includes: a fast charge pump 4202;a peak detecting circuit 4204 having a diode 4206 and a resistor 4208; acapacitor 4210; a resistor 4212; and a comparator 4216. The fast chargepump 4202 receives the RF signal 3003 and detects the amplitude envelopeof RF signal 3003, to generate a demodulated output 4203. For example,FIG. 43A illustrates an example RF signal 3003 that is amplitudemodulated. The fast charge pump 4202 detects the amplitude envelope 4302of the RF signal 3003, and generates the demodulated output signal 4203that substantially tracks the amplitude envelope 4302 of the RF signal3003. The demodulated output signal 4203 is further processed by thecomparator 4216 to generate a digital output signal 4220 that representsthe demodulated output signal 4203.

The demodulated output signal 4203 is forwarded in parallel to thecomparator 4216 and to peak detector 4204, where the peak detector 4204generates a reference signal 4214 that is used as a threshold voltage bythe comparator 4216. More specifically, the diode 4206 in the peakdetector 4204 conducts when the demodulated output signal 4203 risesabove the voltage at the node 4209, and the resulting charge is storedon the capacitor 4210. This typically occurs when there is a transitionfrom a “0” to a “1” in the demodulated output signal 4203. When there isa transition from a “1” to a “0”, then the demodulated output signal4203 falls below the voltage at the node 4209, and the excess charge atthe node 4209 is bled off through the resistors 4208 and 4212. As aresult, reference voltage 4214 at node 4209 generally follows andapproaches the demodulated output signal 4203, as shown in FIG. 43B, butwith a much longer time constant. The reference signal 4214 if forwardedto the comparator 4216 and operates as a threshold voltage so that thecomparator 4216 can detect data transitions in the demodulated outputsignal 4203. The comparator 4216 compares the amplitude of thedemodulated output signal 4203 with that of the reference signal 4214,and generates digital output signal 4220 that is representative of thecomparison.

The relative sizes of the capacitor 4210, resistors 4208, 4212, and thediode 4206 determine how closely the time constant of the referencevoltage 4214 tracks the demodulated output signal 4203. If the referencevoltage is too responsive, then the comparator 4216 will not be able todetect the desired data transitions (i.e. “0” to “1”, and “1” to “0”) inthe demodulated output signal 4203 because the signals will be too closeto each other. In one embodiment, an input voltage offset is implementedin the comparator 4216 to insure sufficient separation between the twosignals so that a proper comparison can be made. For example, thevoltage offset can be implemented by skewing the sizes of the inputtransistors in the differential inputs of the comparator 4216. Inembodiments of the invention, the voltage offset is set to approximately25 mV so as to insure sufficient separation between the demodulatedoutput signal 4203 and the reference signal 4214.

FIG. 44 illustrates the fast charge pump 4202 according to oneembodiment of the present invention. The fast charge 4202 includes twostages 4402 a and 4402 b, where each stage 4402 has a similar structureas the charge pump 4000, but without the voltage limiting shunt MOSFETdevices to ground, (e.g. without the MOSFET devices 3404 in the chargepump 4000). The shunt MOSFET devices are not needed during datarecovery, since the output 4203 is not being used as a voltage supplyfor other circuits. Furthermore, the voltage clamping effect of thepower charge pump causes excessive RF energy to reflected away from theantenna, which will preclude the generation of a output voltage 4203that is too large.

The fast charge pump 4400 is illustrated as having two stages 4402.However, the fast charge pump is not limited to two stages. The chargepump 4400 can have any number of stages as will be understood by thoseskilled in the relevant arts.

The components in the fast charge pump 4202 are configured so that thefast charge pump 4400 has sufficient bandwidth to track the modulationon the RF signal 3003. For instance, the MOSFET diodes 4406 and 4408 aresized smaller than the corresponding diodes in the power charge pump4000. Smaller sized devices have smaller parasitic reactance andcapacitances. Therefore, smaller sized devices are more responsive tohigher bandwidth signals than that which can be achieved with largerdevices. The capacitors 4404 and 4410 are also appropriately sized toprovide sufficient bandwidth. As a result, the charge pump 4400 hassufficient bandwidth and is fast enough to track the modulation on theRF signal 3003, and generate the demodulated output signal 4203.

Furthermore, the charge pump 4400 includes the efficiency improvementsdiscussed above for the charge pump 4000. Namely, the body terminals ofthe MOSFET diodes 4406 and 4408 are connected to the respective drainterminals of these devices, which improves the steepness of therespective IV curves of the MOSFET diodes 4406 and 4408 through backbias threshold reduction. Second, the gates of the diodes 4406 areforward biased to shift-up the operating point of these diodes on theirrespective IV curves.

As shown in FIG. 44, the fast charge pump 4400 also includes an extraMOSFET diode 4412 that is not needed in the generation of thedemodulated output signal 4203. The MOSFET diode 4412 is the diode 4206in the peak detector 4204, and is used to generate the reference signal4214 that is fed to the comparator 4216. Therefore, the fast charge pump4400 generates both the demodulated output signal 4203 and the referencesignal 4214 that are used by the comparator 4216 for thresholddetection.

Referring back to FIG. 30, the power charge pump 3008 and the datarecovery circuit 3010 are configured in parallel and can operatesimultaneously on the same RF signal 3003. Therefore, voltage generationand data recovery can occur simultaneously, which improves the responsetime of the tag.

To summarize the data recovery circuit, the basic charge pump design isused as a signal detector with the MOSFET diodes and the charge pumpcapacitors all sized approximately an order of magnitude smaller, sothat the charge pump follows rapid changes in the modulated RF signal.Also, the clamping devices are removed to allow the charge pump outputto follow the input signal strength variations of the modulated RF. Inorder for the charge pump to detect signal strength changes, a unique“self-reference” feature is added, by connecting the output through aMOSFET diode to a holding capacitor. The holding capacitor will follow atime integrated signal strength, whereby the charge output will followthe rapid changes of the modulated RF signal. By comparing the two, witha standard comparator circuit, signal detection is achieved.

5.3 Transmit Modulation

FIG. 45 further illustrates the backscatter switch 3016 in FIG. 30. Theback scatter switch 3016 modulates the impedance of the antenna 3002 totransmit data back the reader. The backscatter switch 3016 includes aMOSFET transistor switch 4504 connected to the input of the antenna atthe node 3005. The MOSFET switch 4504 has its drain connected to theinput of the antenna 3002 through a DC blocking capacitor 4502, and itssource connected to ground. The gate of the MOSFET switch 4504 iscontrolled by the transmit data 3017 so that the antenna 3002 is shortedto ground through the capacitor 4502 according the transmit data 3017.More specifically, the backscatter switch 3016 shorts the inputimpedance seen by the antenna 3002 at the RF pad 3004. Therefore, the RFsignal 3003 is reflected back to the reader in accordance with thetransmit data 3017.

6. Conclusion

While various embodiments of the present invention have been describedabove, it should be understood that they have been presented by way ofexample only, and not limitation. Thus, the breadth and scope of thepresent invention should not be limited by any of the above-describedexemplary embodiments, but should be defined only in accordance with thefollowing claims and their equivalents.

1. A method for defining data symbols in a radio frequencyidentification (RFID) tag device, comprising: (a) receiving a firstcalibration signal; (b) detecting a characteristic of the received firstcalibration signal; and (c) storing the detected characteristic of thereceived first calibration signal as a stored first characteristic;wherein a received data symbol is determined to be a first data value ifa characteristic of the received data symbol has a first predeterminedrelationship with the stored first characteristic; and wherein thereceived data symbol is determined to be a second data value if thecharacteristic of the received data symbol has a second predeterminedrelationship with the stored first characteristic.
 2. The method ofclaim 1, wherein step (a) comprises: (1) receiving the first calibrationsignal on a carrier signal.
 3. The method of claim 2, wherein step (a)further comprises: (2) storing charge from the received carrier signalin at least one capacitor to a direct current (DC) voltage, wherein thestored DC voltage is used to power the RFID tag device.
 4. The method ofclaim 3, wherein step (2) comprises: rectifying the carrier signal in adiode rectifier circuit.
 5. The method of claim 4, wherein step (2)comprises: rectifying the carrier signal in a diode rectifier circuithaving a plurality of stages.
 6. The method of claim 5, wherein saidrectifying step comprises: using the plurality of stages to cause the DCvoltage to be greater than a received voltage of the carrier signal. 7.The method of claim 1, wherein the characteristic of the received firstcalibration signal is a frequency of the received first calibrationsignal.
 8. The method of claim 1, wherein the characteristic of thereceived first calibration signal is an amplitude of the received firstcalibration signal.
 9. The method of claim 1, wherein the characteristicof the received first calibration signal is a phase of the receivedfirst calibration signal.
 10. The method of claim 1, wherein thecharacteristic of the received first calibration signal is a length of apulse portion of the received first calibration signal.
 11. The methodof claim 1, wherein the characteristic of the received first calibrationsignal is a length of a cycle of the received first calibration signal.12. A system for defining data symbols in a radio frequencyidentification (RFID) tag device, comprising: means for receiving afirst calibration signal; means for detecting a characteristic of thereceived first calibration signal; and means for storing the detectedcharacteristic of the received first calibration signal as a storedfirst characteristic; means for determining a received data symbol to bea first data value if a characteristic of the received data symbol has afirst predetermined relationship with the stored first characteristic,or to be a second data value if the characteristic of the received datasymbol has a second predetermined relationship with the stored firstcharacteristic.
 13. The system of claim 12, wherein said means forreceiving a first calibration signal comprises: means for receiving thefirst calibration signal on a carrier signal.
 14. The system of claim13, wherein said means for receiving the first calibration signal on acarrier signal comprises: at least one capacitor that stores charge fromthe received carrier signal to produce a direct current (DC) voltage,wherein the DC voltage is used to power the RFID tag device.
 15. Thesystem of claim 14, wherein said means for receiving the firstcalibration signal on a carrier signal further comprises: a dioderectifier circuit that rectifies the carrier signal.
 16. The system ofclaim 15, wherein said diode rectifier circuit has a plurality ofstages.
 17. The system of claim 16, wherein said plurality of stagescause the DC voltage to be greater than a received voltage of thecarrier signal.
 18. A method for communicating with radio frequencyidentification (RFID) tag devices, comprising: (a) transmitting acalibration signal having a characteristic; and (b) transmitting a datasymbol; wherein a RFID tag device that receives the calibration signaland the data symbol uses the characteristic to determine a data value ofthe data symbol.
 19. The method of claim 18, wherein the characteristicof the calibration signal is a length of a pulse portion the calibrationsignal.
 20. The method of claim 19, wherein the RFID tag devicedetermines the data symbol to be a first data value if a length of apulse portion of the data symbol is less than the length of the pulseportion of the calibration signal; and wherein the RFID tag devicedetermines the data symbol to be a second data value if the length ofthe pulse portion of the data symbol is greater than the length of thepulse portion of the calibration signal.
 21. The method of claim 18,wherein the RFID tag device determines the data symbol to be a firstdata value if a characteristic of the data symbol has a firstpredetermined relationship with the characteristic of the calibrationsignal; and wherein the RFID tag device determines the data symbol to bea second data value if the characteristic of the data symbol has asecond predetermined relationship with the characteristic of thecalibration signal.
 22. The method of claim 18, wherein thecharacteristic of the calibration signal is a frequency of thecalibration signal.
 23. The method of claim 18, wherein thecharacteristic of the calibration signal is an amplitude of thecalibration signal.
 24. The method of claim 18, wherein thecharacteristic of the calibration signal is a phase of the calibrationsignal.
 25. The method of claim 18, wherein the characteristic of thecalibration signal is a length of a cycle of the calibration signal. 26.The method of claim 18, wherein step (a) comprises: transmitting thecalibration signal on a carrier signal.
 27. A method for communicatingwith radio frequency identification (RFID) tag devices, comprising: (a)transmitting a first calibration signal having a first characteristic;(b) transmitting a second calibration signal having a secondcharacteristic; and (c) transmitting a data symbol; wherein a RFID tagdevice that receives the first calibration signal, the secondcalibration signal, and the data symbol uses the first and secondcharacteristics to determine a data value of the data symbol.
 28. Themethod of claim 27, wherein the RFID tag device determines the datasymbol to be a first data value if a characteristic of the data symbolhas a first predetermined relationship with the first characteristic;and wherein the RFID tag device determines the data symbol to be asecond data value if the characteristic of the data symbol has a secondpredetermined relationship with the first characteristic and has a thirdpredetermined relationship with the second characteristic.
 29. Themethod of claim 28, wherein the RFID tag device determines the datasymbol to be a third data value if the characteristic of the data symbolhas a fourth predetermined relationship with the second characteristic.30. The method of claim 27, wherein the first characteristic is a lengthof a pulse portion the first calibration signal, and the secondcharacteristic is a length of a pulse portion of the second calibrationsignal.
 31. The method of claim 30, wherein the RFID tag devicedetermines the data symbol to be a first data value if a length of apulse portion of the data symbol is less than the length of the pulseportion of the first calibration signal; wherein the RFID tag devicedetermines the data symbol to be a second data value if the length ofthe pulse portion of the data symbol is greater than the length of thepulse portion of the first calibration signal and less than a length ofthe pulse portion of the second calibration signal; and wherein the RFIDtag device determines the data symbol to be a third data value if thelength of the pulse portion of the data symbol is greater than thelength of the pulse portion of the second calibration signal.
 32. Themethod of claim 27, wherein the first characteristic is a frequency ofthe first calibration signal, and the second characteristic is afrequency of the second calibration signal.
 33. The method of claim 27,wherein the first characteristic is an amplitude of the firstcalibration signal, and the second characteristic is an amplitude of thesecond calibration signal.
 34. The method of claim 27, wherein the firstcharacteristic is a phase of the first calibration signal, and thesecond characteristic is a phase of the second calibration signal. 35.The method of claim 27, wherein the first characteristic is a length ofa cycle of the first calibration signal, and the second characteristicis a length of a cycle of the second calibration signal.
 36. The methodof claim 27, wherein step (a) comprises transmitting the firstcalibration signal on a carrier signal; and wherein step (b) comprisestransmitting the second calibration signal on the carrier signal.
 37. Aradio frequency identification (RFID) reader, comprising: a transmitter;and an antenna coupled to an output of said transmitter; wherein saidtransmitter is capable of outputting a calibration signal fortransmission by said antenna, wherein said calibration signal has acharacteristic, wherein said transmitter is capable of outputting a datasymbol for transmission by said antenna; wherein a RFID tag device thatreceives said calibration signal and said data symbol uses saidcharacteristic to determine a data value of said data symbol.
 38. TheRFID reader of claim 37, wherein said calibration signal is transmittedon a carrier signal.
 39. The RFID reader of claim 38, wherein the RFIDtag device determines said data symbol to be a first data value if alength of a pulse portion of said data symbol is less than said lengthof said pulse portion of said calibration signal; and wherein the RFIDtag device determines said data symbol to be a second data value if saidlength of said pulse portion of said data symbol is greater than saidlength of said pulse portion of said calibration signal.
 40. The RFIDreader of claim 37, wherein the RFID tag device determines said datasymbol to be a first data value if a characteristic of said data symbolhas a first predetermined relationship with said characteristic of saidcalibration signal; and wherein the RFID tag device determines said datasymbol to be a second data value if said characteristic of said datasymbol has a second predetermined relationship with said characteristicof said calibration signal.
 41. The RFID reader of claim 37, whereinsaid characteristic of said calibration signal is a frequency of saidcalibration signal.
 42. The RFID reader of claim 37, wherein saidcharacteristic of said calibration signal is an amplitude of saidcalibration signal.
 43. The RFID reader of claim 37, wherein saidcharacteristic of said calibration signal is a phase of said calibrationsignal.
 44. The RFID reader of claim 37, wherein said characteristic ofsaid calibration signal is a length of a pulse portion said calibrationsignal.
 45. The RFID reader of claim 37, wherein said characteristic ofsaid calibration signal is a length of a cycle of said calibrationsignal.
 46. A radio frequency identification (RFID) reader, comprising:a transmitter; and an antenna coupled to an output of said transmitter;wherein said transmitter is capable of outputting first and secondcalibration signals for transmission by said antenna, wherein said firstcalibration signal has a first characteristic and said secondcalibration signal has a second characteristic; wherein said transmitteris capable of outputting a data symbol for transmission by said antenna;wherein a RFID tag device that receives said first calibration signalsaid second calibration signal, and said data symbol uses said first andsecond characteristics to determine a data value of said data symbol.47. The RFID reader of claim 46, wherein the RFID tag device determinessaid data symbol to be a first data value if a characteristic of saiddata symbol has a first predetermined relationship with said firstcharacteristic; and wherein the RFID tag device determines said datasymbol to be a second data value if said characteristic of said datasymbol has a second predetermined relationship with said firstcharacteristic and has a third predetermined relationship with saidsecond characteristic.
 48. The RFID reader of claim 47, wherein the RFIDtag device determines said data symbol to be a third data value if saidcharacteristic of said data symbol has a fourth predeterminedrelationship-with said second characteristic.
 49. The RFID reader ofclaim 46, wherein said first characteristic is a phase of said firstcalibration signal, and said second characteristic is a phase of saidsecond calibration signal.
 50. The RFID reader of claim 49, wherein theRFID tag device determines said data symbol to be a first data value ifa length of a pulse portion of said data symbol is less than said lengthof said pulse portion of said first calibration signal; wherein the RFIDtag device determines said data symbol to be a second data value if saidlength of said pulse portion of said data symbol is greater than saidlength of said pulse portion of said first calibration signal and lessthan a length of said pulse portion of said second calibration signal;and wherein the RFID tag device determines said data symbol to be athird data value if said length of said pulse portion of said datasymbol is greater than said length of said pulse portion of said secondcalibration signal.
 51. The RFID reader of claim 46, wherein said firstcharacteristic is a frequency of said first calibration signal, and saidsecond characteristic is a frequency of said second calibration signal.52. The RFID reader of claim 46, wherein said first characteristic is anamplitude of said first calibration signal, and said secondcharacteristic is an amplitude of said second calibration signal. 53.The RFID reader of claim 46, wherein said first characteristic is alength of a pulse portion said first calibration signal, and said secondcharacteristic is a length of a pulse portion of said second calibrationsignal.
 54. The RFID reader of claim 46, wherein said firstcharacteristic is a length of a cycle of said first calibration signal,and said second characteristic is a length of a cycle of said secondcalibration signal.
 55. The RFID reader of claim 46, wherein said firstcalibration signal is transmitted on a carrier signal and said secondcalibration signal is transmitted on said carrier signal.
 56. A radiofrequency identification (RFID) communications system, comprising: aRFID reader that includes a transmitter and an antenna coupled to anoutput of said transmitter; and a RFID tag device; wherein saidtransmitter is capable of outputting a calibration signal fortransmission by said antenna, wherein said calibration signal has acharacteristic, wherein said transmitter is capable of outputting a datasymbol for transmission by said antenna; wherein said RFID tag device isconfigured to receive said calibration signal and said data symbol, andis configured to use said characteristic to determine a data value ofsaid data symbol.
 57. The system of claim 56, wherein said antennatransmits the first calibration signal on a carrier signal.
 58. Thesystem of claim 57, wherein said RFID tag device comprises: at least onecapacitor that stores charge from the received carrier signal to producea direct current (DC) voltage, wherein the DC voltage is used to powerthe RFID tag device.
 59. The system of claim 58, wherein said RFID tagdevice further comprises: a diode rectifier circuit that rectifies thecarrier signal.
 60. The system of claim 59, wherein said diode rectifiercircuit has a plurality of stages.
 61. The system of claim 60, whereinsaid plurality of stages cause the DC voltage to be greater than areceived voltage of the carrier signal.
 62. A radio frequencyidentification (RFID) communications system, comprising: a RFID readerthat includes a transmitter and an antenna coupled to an output of saidtransmitter; and a RFID tag device; wherein said transmitter is capableof outputting first and second calibration signals for transmission bysaid antenna, wherein said first calibration signal has a firstcharacteristic and said second calibration signal has a secondcharacteristic, wherein said transmitter is capable of outputting a datasymbol for transmission by said antenna; wherein said RFID tag device isconfigured to receive said first calibration signal, said secondcalibration signal, and said data symbol, and is configured to use saidfirst and second characteristics to determine a data value of said datasymbol.
 63. The system of claim 62, wherein said antenna transmits thefirst and second calibration signals on a carrier signal.
 64. The systemof claim 63, wherein said RFID tag device comprises: at least onecapacitor that stores charge from the received carrier signal to producea direct current (DC) voltage, wherein the DC voltage is used to powerthe RFID tag device.
 65. The system of claim 64, wherein said RFID tagdevice further comprises: a diode rectifier circuit that rectifies thecarrier signal.
 66. The system of claim 65, wherein said diode rectifiercircuit has a plurality of stages.
 67. The system of claim 66, whereinsaid plurality of stages cause the DC voltage to be greater than areceived voltage of the carrier signal.